AM modulation with solid state 2015

I wanted to make an AM modulator to modulate any sine wave between 100kHz 
and 2MHz.

Googling will show a few schematics for RF signal generators virtually all lack sufficient
info to build anything worthwhile. I hope my schematics here give more understanding
for what could be useful workshop test gear.

The schematics here use discrete discrete solid state devices. There is nothing here which
uses factory made IC function generator chips and 1,001 other supporting parts or op-amps.

AM modulation continues to be widely used in electronics since AM radio transmissions
began by about 1920.
Methods here produce milli-watt level RF signals to simulate AM radio transmissions for
testing AM radios etc.

I wanted AM production with :-
1. AM carrier  RF bandwidth 100kHz and 4MHz, -1dB, 65kHz to 4.5MHz, -3dB,
and NOT using any tuned LC networks.
2. Flat response of audio F modulated envelope between 10Hz and 20kHz, -3 dB poles.
3. Less than 3% THD in envelope shape at 97% mod using 1kHz audio sine wave with
audio THD < 0.01% THD.
4. Envelope THD declining linearly with mod %, say 1.5% at 50% mod,  to negligible levels
towards 10% modulation.
5. Freedom from parasitic oscillations and spurious noise.
6. The same modulated envelope for 97% modulation should occur for a wide range of
RF carrier input levels and for a wide range of AF input levels between 10Hz and 20kHz all
without constantly adjusting audio input levels to get good looking waves at 97% mod.
7. Ability for linearizing the non linear modulation process by local detection of AF to be
used for NFB to reduce envelope shape THD.
8. Ability for use as peak voltage detector  with or without out AM modulation.

There are two input signals required to produce an amplitude modulated signal, the input
"carrier", Vc, and modulating signal, Vm.

The frequency of Vc is usually more than twice Vm. It is possible to have Vc = 80Hz and
Vm = 4Hz, and the arrangement of devices will produce an 80Hz tone which varies in
amplitude at a rate of 4Hz. This idea has sometimes been used in guitar amps with tremolo
In this page I apply the idea where Vc is RF over 100kHz and Vm is audio signal between
5.0Hz and 20kHz or slightly higher.

The basic principle of producing AM is :-
1. Active devices are set up with fixed input low level input of Vc, the carrier, in this case RF.
2. Vc carrier current change in active devices is low.
3. Vm, a modulating signal with different F to Fc is applied to cause a higher change of current
change in the active devices, bjts, mosfets, j-fets, or vacuum tubes.
4. The modulating Vm signal causes device Gm to change approximately linearly to applied
Vm voltage, and the Gm can be reduced to zero, or to about twice the idle Gm.
5. The change of Vc carrier current in devices = Vm x Gm. The amplitude of the Vc current is
thus changed between zero and twice the idle value, and this gives maximum possible
amplitude modulation of carrier signal.

The basic process is the same for all Vc frequencies including AF or RF and circuits may be
similar, but C, L and R values chosen may be very different for the range of frequencies
This page explores the use of Vc < 100kHz and change of current and Gm in mosfets
and bjts at modulation F between 5Hz and 20kHz.

The changes in currents of both LF modulating signal and HF fast carrier signal produce signal
voltages in any load resistors which are used in series with drains or collectors.
The RL signal voltage in Vrms = Gain of device x input voltage. The gain of devices with high
internal drain or collector resistance such as mosfets and bjts = Gm x RL, where
RL < R internal / 20. The voltage gain is also subject to effect of device capacitances but we
need to ignore C while considering basic function at fairly low F.
Fig 1. VERY basic AM gene.
Fig 1 Vc carrier input F can be between 10kHz and 2MHz, but here I found operation between
100kHz and 1MHz to be fair. When Vc has its amplitude varied linearly with a lower F audio signal,
the carrier "carries" the audio wave information without the presence of the modulating signal.
Where Vc > 100kHz, it is easily transmitted across air or space as RF energy, and so much
more efficiently that shouting at each other over long distances or using telegraph poles and
wires, although fibre optic cables and satellites do give relief to the mass of messages we send! 

The modulating signal is Vm, and here is considered between 5Hz and 20kHz.
Carrier amplitude changes convey the volume change heard, and the peaks and valleys of envelope
shape conveys the AF frequencies. The envelope shape for music & speech shows constant
complex carrier envelope shape. But with Vm = ONE sine wave at ONE AF, we can see the
Vc envelope shape change as a regular pattern on a CRO and we see that its outline boundaries
above and below the 0V axis have the shape of 2 phases of Vm. 

In Fig 1, input Vc = 100kHz = 2.2Vrms. It is reduced by network C3+R3+R4 to 200mVrms and
applied to Q1 gate, and the signal supply resistance of at both Q1+Q2 gates = 42r
Q1+Q2 work as a simple long tail pair differential amp which produces equal but oppositely phased
Vc signals at drains. Q1+Q2 sources are commonly connected to a high ac impedance current
source of collector of Q3.

Each Q1+Q2 drain has same R load, and carrier Vg-s for Q1 and Q2 is same, but opposite phases
and thus balanced. Q1+Q2 act like any LTP, or differential amp with high resistance current supply
to commoned sources, and with Vc input to only one gate, with other gate grounded. 

The Vc RF level needs to be kept low to minimize THD in Vc. The small  change in carrier RF current
during each carrier wave means mosfet Gm does not change much during each small RF wave.
The change of current at modulation F is much higher so there is a large change of Gm.
The idle current in each mosfet = 4.5mAdc, and this is moved down to 0.0mA and up to a peak 9mA.
The change of current in both mosfets is the same, and has the same phase at modulation F.
The change of current is achieved by connecting both Q1+Q2 sources to collector of Q3 which is a
"linear voltage controlled current source", with Vm audio input voltage controlling current in both mosfets.

With no Vm input, only Vc signals are seen at each drain and with no amplitude change.
Gm of each mosfet in class A with idle Idc = 4.5mA is 16mA/V, so RF gain with RL 120r = 1.92,
so with 0.2Vrms gate to gate Vc input, Vc drain to drain = 0.384Vrms, or about 0.19Vrms at each
drain, and carrier current = 0.19V / 120r = 1.58mArms= +/-2.23mApk, so just under 1/2 the idle
current in each mosfet.

Maximum Vm input is where Id is reduced from idle to 0.0mA, then to a maximum of twice
idle Id, ie,  between 0.0mA and 9mA max. This aims to give 100% AM modulation. You could
further increase Id, but high distortion of envelope occurs. Vm input must be kept within practical
limits. With Vm current change of +/-4.5mApk, Vm signal voltage change at each 120r = 0.0 to 1.08V
=  +/- 0.54Vpk = 0.38Vrms.
Both Vm at both 120r have same phase and appear to have the same wave shape of the audio
Vm input to Q3 base.
Without any Carrier Vc input, but with maximum Vm, only Vm waves are seen at each drain.
But when Vc is applied to gates, you will see Vc appear at drains in addition to a constant Vm.
There is a simple addition of two signals being added to a simple load resistance. But you will notice
the HF Vc has much amplitude change and biggest Vc change is when Vm has reached its maximum
peak level as I show in Fig 2 below.

If Vc input is held constant, and Vm increased above 0V, and Vm and Vc levels are
measured at each
120r, the Vm amplitude change is accompanied by Vc amplitude change of same magnitude, so the
applied Vm is linearly amplitude modulating the Vc.

The process is not perfectly linear. The shape of the modulation envelope can have typical THD = 5%
at 100% modulation when envelope is on the brink of developing flats on wave peaks, ie, clipping.
The envelope THD Where there is 5% THD in envelope shape, it shows in compression of maximum AM
wave peaks, so Vc is being compressed and some 3H or 300kHz is being produced. This distortion
of Vc is seen with a pure R drain load, but would be excluded if drain loading was a tuned LC.

Keeping drain Vc just below 0.2Vrms at idle without modulation gave carrier THD just low enough.

Audio F modulation Vm is applied to to Q3 base. Q3 has high enough emitter load R7 470r so
that it operates as a linear emitter follower with THD in Vm currents < 0.1% at highest wanted levels.
Q3 has far higher Gm than mosfets so its open loop gain is high and local follower NFB ensures a
linear I change with Vm change. The Q3 emitter Re 470r ensures collector resistance is a virtual
pure current source which is linearly controlled by Vm applied to Q3 base.

Q1+Q2 commoned sources form a very non linear input resistance load varying between very high
ohms at 0.0mA and 15.6r at  Ic max 18mA. So the Vm waves at both Q1+Q2 sources look very
distorted and like rectifier signals in a PSU.
Max Vm change at Q1+Q2 sources is about +0.8Vpk and - 0.2Vpk..
Fig 1 shows VR1 pot set up for slightly changing Vdc gate bias to balance idle Idc in Q1+Q2.
This is important to prevent distortions and to equalize amplitudes of outputs at each drain.

This simple AM modulator is not perfect. Some THD of Vc and Vm is produced.
If Vm voltage content at each drain is removed entirely, the "envelope" shape of the modulated Vc
waves may have about  6% THD at 97% modulation. The envelope shape may have mainly 2H
with lower levels of higher H. The envelope THD % reduces at lower levels of modulation
so is low at 10% modulation but rate of increase in THD is not linear, and depends on unpredictable
transfer curves.
If Vc is transmitted with envelope distortion, an AM radio receiver will reproduce all THD in
detected AF signal. Many old radios may add several % of THD and IMD so it easy to hear
the shortcomings limitations of poorly modulated and transmitted and received AM radio.

CRO 1. Views of AM Envelope shape distortion.
CRO 1 shows what range of possible distortion may be found during construction
of a SIMPLE AM modulator.
Most ppl who try to make an AM modulator to give less than 5% envelope THD
at 100% modulation will find it very difficult because there are just too many things
to get just right, such as :-
Definition of mission plan,
Device choice, understanding of basic device properties,
Idle current in devices,
Resistance termination of device electrodes,
Level of applied Vc to input of current modulated devices
Level of applied Vm to modulating devices,
Values of R, L and C.
All the distorted wave forms are a result of one or more things which go not allow
a linear change of device Gm with the applied modulating voltage. The challenge is to
overcome the problems to give envelope THD less than 5% at 99% modulation.  

COWPAT = 1 / N squared.
Chance Of Working Perfectly Any Time = 1 / ( Number of things you did not get right )
If there are 6 things, the COWPAT = 1/36, = 2.77%, so never kid yourself that
you are like the Pope, and Infallible.

To properly test AM radios, an RF gene is used and should produce AM waves to test the
whole AM broadcast band
between 522kHz and 1,719kHz and the IF amp which may need
IF signals between 100kHz and 480kHz, with most common IF = 455kHz.
I have not tried to make AM modulators for F above 2MHz. If the AM detection in AM radios
is found to have low THD at very low levels which do not generate AGC voltage, they may
usually be found to sound OK when detecting short wave bands between say 5MHz to 22MHz.

If IF amp and AF detector are found to work OK then the alignment of SW band LC need
only be for maximum signals over widest spread of F, and to maintain tracking where the
difference between tuned RF input and local oscillator Fo is the IF, often 455kHz. Sadly, this is
not often the case in all AM radios, including many much loved sets dating back to 1930.

The effort on this page is to explain AM modulation with SS, and does not provide details
for any tunable RF oscillator, aka VFO.  

It is easier said than done to make a good modulator as an add-on unit with a handful of
discrete devices and which is usable with most simple stand alone RF and AF generators.

Fig 2. Vc 455kHz + Vm 30.33kHz for Fig 1 schematic
Fig 2 shows working wave shapes in schematic in Fig 1 :-

A = Vm input signal voltage to Q3, 30kHz.

B = Common source Vac at Fig 1 Q1+2 sources, and it is at 30kHz but with very high
even numbered H products because of large change of non linear source input resistance.

C = Waves at ONE of the drain load resistors, say R6, 120r.
At the other load R8 120r, the same wave seems to appear, but only the VM component
is the same phase and amplitude, while Vc has same change in amplitude but it is of
opposite phase.
D = Wave achieved at each drain load when Vm is removed from signal by following
device stages or by use of CT choke with low resistance, or by use of transformer,
which then excludes the passage of any small amount of Vm which may appear across
the primary with CT.

The carrier = 455kHz and modulation is by 30.33kHz, and modulation depth = 100%.

In the real world, in most signal generators, Vm of 30kHz or higher is never used because
the highest audio F = 20kHz, and highest AF for AM broadcasting < 10kHz, ie, F above 9kHz
are rapidly attenuated.
I have used 30kHz to make the waveform drawings more easily understood with each
individual carrier wave seen fairly clearly. In the real world there is no pixilation and you
see smooth curves on the oscilloscope.

Fig 2-D shows the Vc AM wave without presence of Vm which appears at each R6+R8.
D shows a constant Vc followed by Vc changing its amplitude so the positive Vc peaks
describe a sine wave of 30kHz. The negative Vc peaks also describe a 30kHz sine wave,
but produce an opposite phase of 30kHz.

Vc at R6 and R8 = 0.2Vrms = 0.28Vpeak. I show Vm = 0.56Vpeak, and total maximum
Vpeak-peak of Vm + Vc total = 0.56V + 1.12V = 1.68Vpk-pk. This peak-peak Vac change
across 120r produces 14mApk-pk. This pk-pk current change = sum of Vm current change
of 0.0 to 9mA peak plus Vc current change of 4.7mA when Vc has doubled from the idle level.

Fig 2-B shows common source waves at Q1+Q2 sources. There is high THD because the
change of current is from a virtual voltage controlled current generator Q3 collector, and the
common sources of Q1+Q2 form a resistance = 1 / ( 2x Gm ) where Gm is transconductance
of each mosfet which varies during each Vm wave cycle between 0mA/V and 16mA/V.
In other words, the input resistance looking into commoned mosfet sources is highly non
The Gm is approximately proportional to source / emitter current, so Gm varies with Vm input
to Q3, but the action produces about 6% THD in envelope shape even when most variables
are optimized, showing the process is far from being perfectly linear.

The waves in 2-C at each of Q1+Q2 drains look strange, but comprise 2 simply mixed signals
added together, Vc + Vm, with amplitude of Vc varying because the Gm change during each cycle
of Vm and the Vm waves appear linear because the current input change to the sources is linear,
with Vm current THD < 0.1%.

In practical circuits, the output from drains or collectors with R loads at C are subjected to a LPF
which excludes the Vm content. The wanted HF Vc content remains and is shown in wave D.
The LPF function can be achieved by using CT choke feed to Q1+Q2 drains or by a following
differential amp stage with high common mode rejection. Opposite phase differential Vc is unaffected,
but common mode Vm is nearly eliminated.

Complete elimination of Vm is only possible with the choke loading of modulator devices and as
shown in Fig 5 and Fig 6 below for the best SS AM generator I could invent and with good common
mode rejection of Vm.

Fig 3. Shows a 100% modulated Vc at 455kHz with Vm 30.33kHz.
The spectrum analyzer shows there are 3 frequencies present, lower side-band, LSB,
Carrier, and upper side-band, USB.

The analyzer would show four F present at each of
Q1+Q2 drain outputs in Fig 1 because
of presence of Vm.

Fig 4.
Fig 4 shows identical operation to schematic in Fig 1 - but with the addition of L1,
a toroidal cored choke with CT to feed Idc to each mosfet Q1+Q2.

The common mode Vm signals at drains is suppressed because the choke inductance
couples each half winding magnetically and common mode voltage applied to each end
of choke create opposing magnetic fields, and current cannot flow differentially.
However, some Vm current does flow across the choke where current in Q1 and Q2
is not exactly equal, ie, slightly unbalanced. Distortion currents related to Vm may also
appear where you don't want any.
XL1 15mH = 95r at 1kHz, high enough to allow differential Vm signal to appear at drains
but there is very low differential signal and some very low common mode Vm present is
generated in wire resistance of each 1/2 winding of choke.
The 120r drain loads must be used to ensure there is a fixed finite R load for Vc.
Without 120r, differential Vc output levels would vary hugely because of variation of inductive
reactance ohms for Vac applied. Without a low value shunting 120r. L1 becomes a tuned LC
with whatever self C exists in the coil plus C looking into mosfet drains. The Fo resonant
F will be within the desired RF bandwidth, and above this F the total shunt C reactance
causes reactive load value to reduce with higher F. To reduce effects of reactive loading
and resonances, the L1 is shunted by resistance lower than XL or XC within the wanted
bandwidth. The modulator output resistance is determined by 120r, and the circuit is immune
to outside loads of following stages.
With R = 120r, the C for a -3dB change of load and gain drop at say 2MHz is  662pF,
so that a following stage of load of 220pF won't reduce Vc by -3dB until 6MHz.

In practice, getting to 6MHz with Fig 1 circuit is impossible because of other internal
shunt C in mosfets all working together to stop you getting what you imagine might be

I found it impossible to get a nice looking envelope shape at 100% modulation and
at 2MHz, no matter what changes I made to Vm and idle Idc and feed of Vc to gates.
Possibly the use of drain loads as low as 12r0 and gate input loading of 5r6 may overcome
C effects, but then output drain signal levels would be 0.02Vrms instead of 0.2Vrms.
and wanted high Vo levels at output of unit would be much lower.

L1 Core should be equal to that advertised by Jaycar, material "type 15".
It is Cat No LO-1238, a toroid, unpainted, dark grey. It has permeability µ at 20kHz above
10, much higher than a ferrite rod. Jaycar also have a slightly larger yellow painted toroid
which has a lower µ so don't use this.
The grey LO-1238 needs less turns for a given L than a rod or an air cored coil which must
have much larger dia and many more turns with higher wire resistance and higher self

I used about 2 Meters of Cat 5 with blue outer sleeve removed, and wound about 28
turns of the 4 pairs of polythene insulated wires wires to fill the toroid hole fully.
The centre of the 2M wires were anchored with tape at a starting point on core, then 1M
each side of center was easily threaded around through toroid hole until one layer is achieved,
about 14 turns. The other 14 turns are wound over top of first 14 turns to give a total of
28t for each of 8 wires.
The 8 windings were connected in series on a small adjoining terminal board.
Measured L was about 15mH at about 10kHz, XL = 942r, and at 100kHz it is at least 9k0,
so there is negligible inductive load at all F above 100kHz. If total R shunting L = 240r,
the mosfets are loaded only by R at wanted F for Vc. Shunt C in coil plus shunt C between
drain and source prevents response going as high as I might like, so use of RL on each
side of choke could be as low as 33r, which will reduce Vc output from 0.2Vrms to
0.055Vrms, a rather tiny signal, but it probably would allow F extension to 4MHz.

The IRF610 are a popular RF power mosfet but they do have fairly high internal
capacitances so operation with untuned L+C drain loads requires low R load values
to maximize HF bandwidth. You are free to experiment with other mosfets to get better

I found IRF610 gave higher envelope THD than BF469 bjts which roughly equaled
THD produced by a pair of 6EJ7 pentodes with screen modulation.

Fig 5. BJTs used for AM modulator.....
Fig 5 has Q1 as split load phase inverter feeding balanced Vc input to Q2+Q4 bases.
Q3+Q5 form a Darlington follower for a voltage controlled current source to alter
emitter current of Q2+Q4.
The Vc input needed is much less for BF469 because the bjt gm is 10 times higher
than IRF610, and to produce the same signal at collectors the base Vc signal may
be much lower. Q1 provides good HF performance to 7MHz.

L1 choke with CT feeds Idc = 4.6mAdc to each Q2+Q4 collectors. The pair of BF469
were chosen from 20 samples to have hfe = 135 +/- 5%. R&C emitter bias networks
ensure Idc balance.

The linearity of Q3+Q5 as a follower deliberately reduced slightly by using a low
value of emitter resistance R19, only 33r. Only 0.22Vrms is required across R19
for current change of +/- 9.2mApeak. The Q3+Q5 open loop gain is less than 12,
and the 2H generated is about 3% at max Vo. This has opposite phase to envelope
2H so the THD of Q3+Q5 is reduced about 1/2 by cancellation during modulation.

R15+R17 Loads on Q2+Q4 collectors = 120r, and following stage bias R = 1k2.
The load values may have to be reduced to give a total of 50r if that is the only way
to get a flat RF response to 5MHz.
Notice R23 820k which has no effect on signal function but pulls the negative sides of
C6+C9 towards 0V to polarize the electro C properly.
Examining envelope linearity.

There are several ways to check if modulation is linear.
1. The simplest way uses triangular Vm input wave at 1kHz, with and compare the
Vm with Vc modulated wave shape with a dual trace CRO. It is easy to overlay the
Vm on top edges of envelope shape and adjust Vm to be equal to envelope shape
between 0% up to 50% modulation. The input Vm wave does not have to be a perfect
triangular wave shape because we only want to know the difference in shape between
the Vm and Vc envelope shape. When modulation is increased beyond 50% to 100%,
the increase of envelope shape should remain equal to Vm wave, but this seldom
occurs and envelope increase falls behind the Vm wave by as much as 10% which
is easily seen. The use of sine wave Vm will also give the same result. If envelope
THD is under 2%, it becomes more difficult to see in the CRO using this simple

2. The other classic way to examine modulation linearity is to use a CRO
in X-Y mode with Vm vs Vc. Vm is usually 400Hz, sine wave. Without any modulation,
you see vertical a straight line, but as modulation is increased above 0% you see a
trapezoid appear and at 100% modulation the two sloping sides should extend to a
point leaving a green triangle with 3 nice straight sides.
The two sloped sides of trapezoid should be straight, often the sides curve inward
to the point of 100% modulation which shows the Vc amplitude is failing to remain
linear to amplitude of Vm. But 1% distortion may be difficult to see while 6%+
is very easily seen.
CRO 2. Views of Linearity check.
3. Another method uses dual trace display of envelope and Vm with Vm
overlaid on envelope at 100% mod as mentioned in method 1 above.
The envelope shape of Vc at 100% is adjusted to occupy the full height of CRO
screen on channel 1, and Vm input wave occupies 1/2 height on Channel 2.
In Dual channel mode, Vm is overlaid on top or bottom 1/2 of envelope depending
on phase of Vm. Then you press the "add" button where the CRO adds the Vm
to Vc or you can press the "diff" button where CRO displays difference between
Vc and  Vm. Either way you will see ONE modulated Vc envelope for total height
of screen, and a flat boundary line at bottom or top of screen. This is the result
of simply mixing Vc with Vm.

Some CRO adjustment of Vm amplitude will remove slight ripples in flat boundary
of Vc. Some ripple in this line has same F as the Vm, but cannot be fully reduced
because of phase difference between Vc envelope and input Vm. Vm F can usually
be changed between say 100Hz 2kHz to find F where there is minimum phase shift
and Vm riple can be nulled fully. The remaining ripple in flat boundary of Vc will be
higher F than Vm, mostly 2H or 3H, which is the THD in the envelope, but not in Vm.
But when envelope THD < 0.5% with fairly modulation at 100% mod, the Vc flat
boundary has imperceptible ripple.

When you examine a few AM modulators the linearity differences become very
easy to see.

The THD % = 100% x ( pk-pk ripple / pk-pk envelope shape. )

Relative pk=pk voltage levels are easily counted using the CRO screen divisions.
My Hitachi CRO has 8 main divisions across and down, like a chess board with
64 squares, and each side of square is divided into 5 "graticules" .

A typical result is flat boundary envelope ripple = 2 graticles pk-pk, total envelope
= 40 graticles, so envelope THD = 100% x 2 / 40 = 5%.

When THD is reduced below 1% at 97% modulation, you will have trouble
seeing it on the CRO, and you then have done a fairly good job!

CRO 3. Views of AM Vc wave minus Vm modulation wave

4. The other method involves use of low THD Vm input sine wave at
say 1.0kHz. The Vc AM wave is detected using a low THD envelope detector.
Most simple diode+C+R detectors can get their THD down to 1% at 100%
mod, so if you measure 2% THD in detected AM at 97% mod then you have
about 1% of THD added by modulation process. It is possible to make an
envelope detector giving much less THD, so that if Vm input THD = 0.001%,
and detector THD = 0.1%, and you measure 1%, then the modulation process
is adding nearly 1% THD.
AM modulators are prone to additional oscillations higher than Fc and
appear as wave forms
riding on the carrier. When viewing envelope shapes of Vc waves on a CRO,
you may see bands of faint changes of green intensity in displayed envelope
shape. This indicate the presence of unwanted oscillations. When no spuriae
are present, the green envelope should have uniform green color. Brightening
of boundaries on envelope edges can mean the carrier waves have compressed
appearance, ie, slight flattening of peaks, which indicates generation of 3H.
There is quite a lot to see on a CRO if you know what to look for. A good
radiologist will see much more on an X-ray than you might see.

Fig 6. RF amp, source follower output and choke, switched attenuator.
Fig 6 shows a balanced RF amp has gain of about 12x using 2 x BF472, pnp, with
local current FB and low RL values and high idle current to keep RF response flat to
above 6MHz. This stage follows Fig 5 modulator schematic.

Output from Q6+Q7 collectors directly drive gates of Q8+Q9 IRF610 in source
follower mode which drive RFT1 ( auto-transformer ) with CT which gives 4 output taps
each side for 4 switched output levels.
RFT1 has 8 windings using 4 pairs of lightly twisted wires from Cat-5 cable. It can be
the same as
described above, but below Fig 4. The 8 windings are symmetrically in series
so connections of ends of windings form output taps taken to a 2 pole x 4 pos rotary
wafer switch to give 4 levels of RF output.

Total voltage gain of Q6+Q7 and Q8+Q9 for 100kHz to 6MHz = approx 12x, so if
input to Q6+Q7bases Vb-b = 0.4Vrms then Vout across whole of L2 choke = 4.8Vrms.
Output resistance at each output for highest Vo = 7r0. For switch positions giving lower
Vo, Rout reduces due to transformer impedance change.

I show Q8+Q9 = n-type IRF610 but BF469 should be just as good, and with
R11+R13 both 75r, idle Idc will be 50mAdc each. Max theoretical Iac = +/-50mApk =
37mArms in each Q8+Q9. If each Vo at each source = 2.4Vrms at highest switch
position, minimum balanced load = 130r, or 65r at both outputs, and for class A
operation. If Vo is drawn from one output only, lowest RL = 33r for 2.4Vrms.
175mW is available. For the lowest switch position with Vo from one output side
at 0.6Vrms, L2 TR = 4:1, ZR = 16:1, so balanced load could be 8r0, or 4r0 from
one output.  RFT1 has good magnetic coupling because all its 8 windings are
effectively multi-filar wound close and all magnetically parallel to each other.
Low loads below the standard 50r for a signal gene RF output can be tolerated.

It is remarkable
that in 1950s HP achieved a good result in their HP606A
which had 6 well calibrated RF ranges for 150kHz to 65MHz and good AM
envelope shape with 2 x 6CL6 pentodes using a 6B4 triode for cathode
modulation. But they had exquisitely crafted coils and 4 gang tuning cap
for the PP RF oscillator with 12AT7 and output tank circuit for 6CL6.
Linearity was made acceptable using DC and AC global NFB.
HP produced
the HP606A with only tubes, then the HP606B with solid state in PSU, but
same RF and AF signal circuitry as 6060A with tubes. It does seem possible
all tubes in HP606A could be replaced with solid state, but you have an
enormous amount of work to get all going as well as HP could achieve.

Conclusions about solid state modulator.

The BF469 bjts in modulator stage worked better than mosfets such as IRF610.
But both device types produced high THD ( over 5% ) in envelope shape at
modulation > 95% with carrier F above 1MHz.

I spent 2 weeks full time trying to optimize the voltage and current settings,
but I concluded these SS devices did not work with untuned loads as well as
vacuum tubes with tuned loads for modulators.

I spent another 2 weeks trying to make a simple AF detector to produce an
AF signal with low THD during detection. This wasn't too difficult. The detector
was to provide AF signal for use as NFB applied to a simple wide band
differential amp so that envelope THD could be reduced from 6% at 90% mod
to less than 1%.

This seemed like such a simple idea which would work, but I soon found
I could not succeed because the smallest application of NFB to correct
envelope shape THD always gave every possible type of parasitic oscillation
and some unusual inexplicable behaviour.

I then concluded that NFB probably was never ever going to work because
the modulator stage has very wide bandwidth and gain for all F between 50kHz
to 3MHz. The AF amp could not rely on gain shelving and phase shift reduction
below 20Hz and above 10kHz to avoid parasitics, similar to how stability in
pure audio amps is achieved.  There seemed to always be an F where
oscillation would occur, or the CRO screen would fill with amplified noise.

The only way I think I could could avoid parasitic F was to use tuned LC in
collector circuits with Q at least 10, so that high gain is confined to the tuned
bandwidth of LC.

I wanted minimum RF to be 100kHz, not that far above the audio band. This is
a bit low for for LC, but I could have I have tried using dual radio tuning gangs,
with each gang giving total C between 48pF to 480pF. These could be arranged
for 3 switched RF bands, 170kHz to 544kHz, and 530kHz to 1.7MHz and 1.7MHz
to 5.4MHz. 3 switched coils with CT are needed, 3.8mH, 0.38mH, 0.038mH.
The complexity required with adding switched coils and tuning C become extremely
difficult to get working correctly with equal Vo levels for each F along each band
and equal for all 3 bands. Then the input RF signal has to come from a suitable
RF oscillator which should ideally have 3 switched RF bands with identical F to
modulator so good tracking must be achieved. The RF output from RF oscillator
must have a fixed output level for all F between 170kHz and 5.4MHz, and set for
optimal operation. All this is ALREADY done in a HP606A tubed RF gene which
has 6 switched RF bands between 150khz and 65MHz, all nicely calibrated.
It soon will become obvious that 3 switched RF bands to 5.4MHz will not be easy
with an ordinary 3 position rotary switch wafer switch because of  unwanted
couplings and oscillations. I have built enough RF genies to know this, so its better
to restore a HP606A or B than build your own modulator.

A PP modulator stage is not essential. SE circuits can work fairly well and this
means all LC can be ordinary unbalanced LC with one L and one tuning gang.

I concluded I could not succeed well enough with solid state and untuned loads,
and I changed direction to re-building a tubed RF gene and modulator with SE
oscillator, and PP modulator needing the same single L in oscillator and output
amp and with a 3 gang tuning C.

Fig 7.
Fig 7 has an op-amp design I have used before for wide bandwidth test gear.
Its application here is based on an idea mentioned by Mr David W Knight of UK
in his online published hand drawn schematics.

Fig 7 is quite simple compared to what would be found in many IC op-amps.
Unlike most IC op-amps, TWO additional outputs are available from collector
current sensing resistors, apart from the usual single terminal for signal output.

The op-amp is capable of 0.71Vrms input without THD up to 5MHz. This is
much better than many op-amps which cannot be used above 200Hz.
When Vc carrier input = 0.707Vrms, and applied to +non inverting input port
of amp it produces about 0.702Vrms across load of R20 100r, a permanently
connected RL. The output to this load is all fed back to input inverting port for
NFB to make the amp work like a normal op-amp in unity gain follower mode.
The load is a pure 100r, with no shunt C so amp is immune from phase shift
with outside world C reactance loading. There is a critical damping Zobel
network R10 100r + C6 100p to ensure there are no parasitic HF oscillations.

Now Q7+Q8 are biased for class B with Idle dc < 1mA. The open loop
voltage gain at say 1MHz is at least 100, and when 5mV input is applied,
Q7 or Q8 base voltage swings +/-0.5Vpk before any R20 load voltage begins
to appear. With more than 5mV input, the Vo at R20 begins to closely follow
the input Vc increase, and R20 load signal becomes linear to Vin because of
the NFB loop. So Vo is virtually equal to Vin, ie, if Vin = 0.707Vrms,
Vo = 0.702Vrms.
Thus the non linear turn on character of the bjts is corrected by NFB, ie, the
crossover THD in output load is reduced to negligible levels. The bjt turn on
curve is like a silicon diode, and this non-linearity is avoided in the output.

The current flow in R20 100r load is then linear to Vin, and each 1/2 wave
of carrier Vin generates linear half wave currents in R16 + R19. If maximum
Vrms across 100r = 0.702Vrms, Vpk = 1.0Vpk, so max current = 10mApk.
R16 & R19 are each 330r, so peak Vac = 3.3Vpk. The Vac at R16 are half
sine waves going negative blow the B+ rail, and at R19 the 1/2 sine waves
are positive going from B- rail.

At 100% modulation, carrier Vc input changes between 0.0V and 2Vpk, so
peak Vac across R16 + R19 vary between 0.0V and 6.6Vpk. These 1/2
wave peaks can only be seen clearly on a CRO when there is no C across
330r, and no following low pass RC filtering to remove carrier half waves
while allowing modulation waves to pass.

Now Q7 & Q8 have very high internal collector resistance and they act like
a voltage controlled current sources to feed Iac to the relatively very low value
collector RL of 330r. So the equivalent circuit is a voltage generator with series
R = 20k approx feeding 330r. The effect of any bypass C across 330r on bjt
current production is negligible.
To see what is happening.........

Fig 8. Half wave average RF voltage detector.
If collectors are shunted to 0V by a large value C, then no Vac appears
across 330r. But the ac current signal is that of a 1/2 wave rectifier and
there is a Vdc change across 330r which is the "average Vdc" level for
1/2 sine waves. If the C is large enough, the change to average Vdc is
slowed right down and the Vdc remains constant regardless of whether
the modulation is anywhere between 0% and 100%. The effect of shunting
collector loads 330r has zero effect on the working of the amp and makes
no change to the current change in bjts unless the amp is driven by such
a high level Vin that Vac across 100r + 330r reduces Vc-e to less than a
few volts, and devices are saturated, because their effective Vdc between
emitter and collector limits voltage and current movement. Normal
operation here is well short of levels causing device saturation or overload.

If we measure the Vdc across a large C across 330r, we will Vdc across
C = average Vdc = 0.315 x average Vpeak Peak. At all levels of modulation %,
the average carrier peak Vac remains constant so that with or without any
C across 330r, a meter with high Z input will read Vdc determined solely by
carrier level. So in this case, average Vdc across 330r varies between 0V
when bjts has virtually zero current, and 3.3Vpk for average peak carrier level.
Vdc across 330r = 0.315 x 3.3Vpk = 1.04Vdc. The same sort of waves
occur in 1/2 wave mains rectifier circuits. If there was a full wave rectifier
of sine waves, average Vdc = 0.63 x average peak levels.

The constants of 0.315 or 0.63 apply only for pure sine waves.
The constants are lower for triangle waves and 0.5 and 1.0 for square waves.
But where carrier input is most often an RF sine wave with THD < 5%,
average voltage constants are correct "enough".

Now let us suppose we reduce the C value from a high value to an
intermediate value so that the RF half waves are shunted, but lower F audio
waves are passed. Fig 6 shows C10 and C11 = 15nF shunting R16 and
R19 330r. The -3dB pole is 32kHz. What the CRO will show across each
330r is much reduced level of 455kHz 1/2 waves, while 10kHz audio
modulation waves will pass without much attenuation. But it all looks like a
horrible mess, so additional RC filtering can be used with two cascaded RC
filters using 4k7+1n0 each giving poles at 34kHz. The end result gives audio
signal with bandwidth > 10kHz, -3dB and very little 455khz ripple content.
There is no AF THD caused by diode forward voltage drop and varying
ripple voltage levels seen in ordinary diode+C+R type of detectors.
Such a detector works well with Vc carrier input < 25mVrms.

You can expect that where you have you have 0.71Vrms carrier input and
100% modulation, you should get two opposite phased audio output signals
each = 0.7Vrms.

In SS AM radio receivers the above detector will work well; they may not
generate IF signals exceeding 1Vpk.
In tubed AM radios, IF signals can be between 4Vpk and 20Vpk from local
stations and signals would overload the Fig 7 schematic above.

For tube radio detectors see my detector schematic using cathode
follower to power diode+C+R detector at Kitchen Radio.

How to build stuff :-

Find huge amounts of time for stuff which may take a year.
Prepare for Despair and Disappointment over poor initial results.

To get good usable results, the following approach is required :-
1. Learn by doing, build small schematics and work upwards.
2. Write down wanted F ranges and expected behaviour during use,
compare ideas with other manufactured test gear, get pencils, erasers,
several 128page A4 exercise books to draw all proposed ideas, and keep
drawing results and schematics and conclusions and test results, every
time you are at your bench. 
3. Read old text books used for university electronics degrees in 1960.
4. Build a 1960 circuit, and wear out a pocket calculator getting all R,
C and L values right.
5. Get it working at one frequency.
6. Ask 101 questions about why it works and how it works, and answer
all questions.
7. Extend F range.
8. Possible help may come using Spice simulation software, but guys in
1960 did have a brain inside head and this isn't rocket science.
9. Eliminate shortcomings of each stage before proceeding to next.
10. Apply whatever measures are needed for wanted correct operation,
ie, low THD across each F band, flat output voltage across each F band.
11. Tidy up untidy circuitry.
12. Make dial plates, calibrate dials and label all knobs and switched
and terminals.
13. Draw up schematic with ALL details and print out copy to always
be placed inside metal box of the test gear to allow instant service ease
without searching forgotten details you worked out 4 years before.
To Education and DIY directory

To Index