Wien bridge oscillator with tubes, March 2015.
Last update of this page, 26 March 2015.
The only schematic difference to earlier versions of this page is for better HF
stability in amp in Fig 2, by adding S1c, and C4a, 2n7, etc. Please use the
Fig 2 schematic as I have it if you copy the design. Just remember, if you make
smallest change to anything I have done, it may result in degraded performance.
Be prepared to solve all your own problems.
Many past sine wave oscillators with tubes had insufficient bandwidth with high THD.
Here are details of two I built which were much better than Hewlett-Packard's
1930s first Wien bridge AF oscillator.
Many old variable F tubed oscillators used LC circuits which are not covered in this page.
I have used tubes to produce the sine waves with both Wien bridge oscillators here.
I begin with a re-engineered 1950s AWA telephone tech oscillator for sine waves
only from 1Hz to 220kHz.
The second has tubes to make sine waves from 1Hz to 2.2MHz. It has a solid
state Schmitt Trigger circuit to convert sine waves to square waves, and has
a solid state output buffer and multi level switched output attenuator. Overall
performance is far better than many units using common audio op-amps.
Picture 1. Re-engineered 1955 AWA oscillator.
Fig 1. Schematic WB tubed oscillator, 2005. 1Hz to 220kHz.
Table 1. Measured R values for WB RC network.
Schematic, WB tubed oscillator, Sheet 1, amp, 1Hz to 2MHz 2013.
Sheet 2. Schematic, buffer output for WB tubed oscillator, Sheet 2.
Sheet 3. Schematic, Schmitt trigger and amp for WB tubed oscillator, Sheet 3.
Graph 1. Square wave harmonic content.
sheet 4. Schematic, Power supply for WB tubed oscillator, Sheet 4.
Fig 2. Small signal bjt discrete bjt op-amp.
Picture 1. Old AWA oscillator sold to PMG ( Govt post & telephone company ) from 1950s.
In around 1995 I was given a tubed Wien bridge oscillator made by AWA Australia
for telephone technicians. Like many ancient oscillators made after about 1933
there were only 3 frequency ranges; this one 20-200Hz, 200-2kHz and 2kHz-20kHz.
The original AWA unit used a dual gang AM radio radio tuning cap to vary F
by switching 3 pairs of resistors with highest R value = 22M, lowest value 220k.
It was very
inadequate test gear for testing hi-fi amplifier amplifier response needing
examination between 1Hz and 220kHz at least.
I recall it had a 6SL7 and 6SN7 used as output tube driving 20k : 600r OPT which
I still have for odd purposes.
I completely re-built
5 frequency ranges, 1Hz-22Hz, 10Hz-220Hz, 100Hz-2.2kHz,
F control uses available Alps dual
gang log 50k "Black" pot with 27mm square
body to alter the F. There are 5 pairs of fixed capacitors switched by
2 pole x 6 position old wafer switch. One position is wasted.
The Alps pot has well matched tracks with continuous wiper contact so that wave
form amplitude "bounce" is minimized. I found each track measured 48.2k max.
With the added series R of 2k2, max R = 50.4k, min R = 2k2, and at centre
rotation ( dial = 12 o'clock ), total R = 11.14k, and pot R = 8.94k which is
0.18 x max pot R.
Some "log pots" produce -20dB attenuation from wiper at the centre position or
"12 o'clock" position, ie, there is 10:1 signal voltage reduction at centre position.
AFAIK this is regarded as a true log pot but depends on source signal
R < 100 x total track R, and assumes the loading of wiper output > 10 x track R.
A true log pot of 50k driven by a cathode follower and for gain pot to a preamp
grid and set at centre has 5k from wiper to 0V and 45k from wiper to source.
( The output resistance from wiper = 5k // 45k = 4k5 ).
Many so called log pots including the Alps I have used have a voltage step down ratio
of approximately 5:1. The "50k" Alps pot I used had track R = 49.6k and between
wiper and 0V track was 8.9k when in centre position.
WB oscillators using pots generally have one end of track connected to the wiper
to make a simple variable R. The Alps gave me a pot range 49k to 0k0. There is a
series R1&R4 = 2k2 each so that each R in Wien RC network is varied between
52.2k max to 2k2 min. This allows F variation between 10 and 220.
At near centre of rotation R = 11.1k and F = 50. F between 10 and 50 fill 1.2 the dial
and numbers 50 to 220 fill the other half so numbers are spread out to allow easy
reading to at least 2 significant figures keeping reading errors less than +/-2%.
The dial readings have been verified by measurement of R in Table 1 below.
With the Alps pot I used, and with a series 2k2 I measured the total R values :-
Table 1. Measured R values with dial pointer set to calibrated dial numbers.
The "50k" pot plus 2k2 gives variation slightly more than max & min values of table
so the real F range is from about 9.8 to 223 and allows for trimming C values.
The pot for a Wien bridge oscillator needs to have a smooth R change giving centre
position attenuation ratio of between 4:1 and 10:1.
This applies where you want each F range to be a decade of F, say 10Hz to 100Hz.
Don't try to use a linear pot. There is no way to add additional R to make a "quasi log"
pot for this application. I tried, nothing works. If a linear pot is used for a decade of F
between 10 and 100, then centre position gives 18.1, and numbers between 18 and 100
become very crammed and the high end of dial.
Few cheap dual gang "log pots" give a smooth R change vs rotation. The $4 types
from Jaycar are mostly not worth buying for a WB oscillator. Some seem linear
for part of the turn, then suddenly change to a lesser R, and maybe again, so the
pot does horrible things to how the dial numbers appear. The two tracks are often
poorly matched and added trimming R do not much improve the pot function.
Cheap pots are unsealed, allowing pollution, dust, and soldering fumes to settle on
tracks pots tend to have loosely riveted track connections.
If you must use such crap, spray insides with WD40, press up rivets more tightly
with long nose pliers, seal around open holes with cardboard and silicone before
soldering. Take your Fluke DMM to the shop and sort through their bin for the
best match of tracks and obtain the highest ratio for track R for each side
of the track with wiper set to centre position. Seek the "most logarithmic"
with highest attenuation ratio at centre you can find. And don't be surprised when
you find they ain't very good when you use them.
I tested an ancient 1dual gang 5.0k Colvern wire wound pot formerly used in WB
oscillator. These were very well made with precise matching and smooth R change.
But they suffer from rust on tracks and one gang had a deep rust spot and could not
be cleaned so could not be used. Other track is like new. In centre position R ratio
is 10k2 : 4k8 which is attenuation ration near 3:1. It is not much better than a linear
pot with 2:1 atten ratio ( centre position ).
For the Colvern used for F between 10 and 100 the fixed R = 1.67k for R min
and 16.67k for R max. At the centre of rotation R = 4k8 + 1.67k = 6k5. The F will
be 25.6, so 1/2 of dial is 10 to 25.6, and other 1/2 is 25.6 to 100, with crammed
numbers towards the 100. People accepted this. The wire wound pot does cause
because wiper "bumps" over individual wire turns, but once adjusted they are
extremely reliable, ( if not rusty ).
The Colverns were "REVERSE LOG" which meant the low F numbers appear
on left side of dial, and as you turn knob clockwise, F rises. This is opposite
to all common log "volume" pots, with increasing signal with clockwise turn.
Probably, the best pot will give double the F number for each 90 degrees of turn.
So at zero turn on right side of dial, F = 10, then +90d gives F = 20, and +180d gives
40, and +270d gives 80, and the remaining 30d gives 100.
Many ready made WB oscillators you might buy or build from a kit have unacceptable
calibration errors because the supplied dial numbers do not match the poor quality
So, if you have an existing variable F oscillator which you wish to restore for use
building amps, try checking calibration with a digital F meter, and be prepared
to change the pot to better brand and re-make the dial.
In other Wien bridge oscillators I built I used a 12 position switch to give 12 F
per decade with F spread apart similarly to standard R values between 1 and 10.
Using a dual gang 24 position switch would be much better, but be prepared to
spend a day or two calculating R values, then using paralleled standard values to
get within 1% of the calculated value.
Wien Bridge oscillators, F = 159,000 / ( R x C ),
159,000 is a constant = 1,000,000 / pye, F is output
frequency in Hertz,
R in ohms, C in uF. BTW, pye = constant = 22/7 = 3.14286.
If you calculate the theoretical F for each chosen R and C value, the measured
F will always be slightly different to the theoretical F because of stray C in the
circuitry and slight differences in R values and R tolerances, and R measurements.
It takes considerable experimentation, adjustment, and patience with final R&C values
before the pointer on the dial gives you the exact frequency indicated, +/- 2%, AND
you find the same dial numbers suit all selected F ranges. But you should find a
a strong correlation between theoretical R, C and F to what is measured.
For audio F response testing we do not need more than 2 significant figures for
F numbers. If we adjust the oscillator pointer to 50Hz on the dial, the exact F could be
between 49Hz and 51Hz and we do not need to have 50.000Hz. The +/-1Hz for 50Hz
setting is within +/- 2% accuracy and is quite OK for 99% of response testing of all that
anyone does with Hi-Fi gear including testing a L+C eq filter network for a loudspeaker.
If you need more F accuracy, use a digital F meter to monitor the oscillator which
may be adjusted to be more accurate, ie, say indicate the high Q peak of a
LF loudspeaker box resonance.
To calibrate my dials :-
1. Cut dial card from good white cardboard to slightly over size shape and size
2. temporarily attach dial card to unit panel and mark in pencil where centre
are on panel, to allow removal and replacement after calibration.
3. Use digital F meter to move dial pointer to exact F numbers and draw lines and
F numbers in light HB pencil.
4. When happy with the F readings for 5 F ranges, remove card from panel and
overwrite pencil with black ink and erase pencil.
5. Trim size of dial card to wanted exact size but without losing positioning lines
6. Brush on semi gloss varnish to both sides of cardboard until saturated when it
should be slightly opaque and cream colored.
7. The dial card may be left to dry and later glued to panel.
8. Alternatively, the panel is sanded where card sits and then painted with varnish
when dial is varnished. Oscillator is tuned on bench to have panel horizontal and dial
placed in correct position and all left alone until next day when varnish has
This is Primitive DIY, but it works OK and you don't need a PC.
The digital meter was also used to trim C values shown at S1A and S1B, so only
one scale was needed for all 5 F ranges.
The digital meter I have is a kit I built and able to measure between 1Hz and 50MHz
in 3 F ranges. It is fine above 20Hz, but below 20Hz it becomes slow to give a reading.
Below 10Hz it cannot give fractions of a Hz, and below 5Hz its unreliable, and very
vague below 2Hz, so to calibrate a dial you need a stop watch and then count the
cycles using a CRO. If you count 12 peaks in 10 seconds, you have 1.2Hz.
The Fig 1 schematic can operate to give a higher F up to 800kHz. But above about
200kHz the dial readings used for lower F ranges become incorrect. Vo level declines,
and you get parasitic F. The tube Miller C, stray circuit C, and grid current will all too
easily affect operation too much. With R values I show, the value of Ce to get to
2.2Mhz is a theoretical 31pF. I found that about 220pF was the smallest C value
I could use before the overall function became awful. If I wanted higher F, a
much bigger change to circuit was needed so I settled for 220kHz.
Feel free to do better with neat PC controlled laser cut dials or machine engraved
or laser printed dials on opaque plastic and with LED lamps behind.
I'm always too busy to get around to finer DIY work which does nothing to improve
the work I do with amps I build. Don't be a tradesman who blames his tools.
Most pots have 300 degrees of turn and I found no need for a reduction drive
so setting the F is fine enough even with 10-220 F numbers instead of the more
common 20-200 or 10-100 F range on dial.
The use of a
pot to tune F does generate a slight amount of Vo level bounce
because of the very slight noise during movement of
wiper on carbon track.
The noise comprise intermittent transient voltages at output thus feeding noise
voltage to NFB so that lamps over-heat slightly, increase ß, reducing Vo, then
lamps cool, and ß reduces, Vo goes higher, and amplitude becomes unstable
and "bounces" a bit after noise stops, because of the lamp's slow thermal time
constant. Many old oscillators had huge bounce problems. I found the zener
diodes Vd2 to Zd5 across R11 fixed the bounce problem.
The Fig 1 schematic is not prone to LF or HF parasitic oscillations where you get
your wanted F but you also get an additional oscillation signal making the unit
unusable. The WB oscillator depends on having a high gain amp with both a
PFB network plus a variable ß NFB network used to control gain.
All such circuits are prone to instability if there is excessive open loop phase
shift in the amp, and PSU rails are not well stabilized.
network has 12 seriesed 12V x 50mA lamps. Each lamp is known as
"grain of wheat" incandescent globe used in model trains etc. With 12V
I = 50mA, R = 240r and power is 0.6Watts with hot bright filaments.
When cold, lamp resistance is approximately 27r.
lamp resistance for large
R change for a small V change is about twice the cold R or about 50r.
To double the cold resistance you need less than 0.5Vac applied and there is a
high rate of R change for a given small increase of voltage.
The 12 lamps should be fairly well matched or else one or more will tend to turn
on more than others, so that effective lamp R is not what you want it to be.
There are very few if any available lamps with high R per Volt and which are able
to be used with output tubes such as 6V6, 6F6 in old WB oscillators.
Thermistors are possible in theory in the NFB network but AFAIK, nothing suitable
is available. The best were vacuum sealed inside a glass tube with delicate
element for use on input side of NFB network. The delicate element gives high
amount of R easily driven by tube or transistor and gives high rate of R change for
small voltage change. The glass helps stabilize temperature, and stops heating from
surrounding hot things.
But I don't know if such thermistors are now available, I could not find a source.
The thermistors have lower R as they heat up, so when Vo rises the ß increases
and NFB increases, gain reduces and Vo is kept constant.
The lamps give low enough THD and work better than many other ideas since 1930s
when they were first used in commercially made WB oscillators. Many special things
made before 1970 for odd uses such as control elements in PFB or NFB circuits
are no longer made.
The adjustment of VR3 10k pot adjusts the gain and Vo slightly, and you may find
that as you adjust this pot for least THD the Vo tends to become more "bouncy"
and unstable. While it might be possible to get much less THD than I have here
the price to be paid is instability, so its better to settle for THD just under 0.5%
at 1kHz, than have bounce problems when THD = 0.1% or less.
I have arranged an output stage of amp to be 6AC7 pentode to provide high gain
to a White follower buffer using two 6CM5 in triode, each with Idc = 25mA.
The 6AC7 is a metal can tube from 1940s and I had a lot of them. They have
quite high gM compared to say 6SJ7 and derivatives. 6SH7 might be usable.
A mini nine pin 6BX6/EF80 could also be used.
The White follower enables loading by both PFB and NFB networks without much
affecting amp gain.
Total RL minimum = 1k3 with NFB and PFB in parallel at HF end of the F ranges.
The White follower produces 16.5Vrms and this is fed to the output attenuator pots
for two output voltage ranges, up to 14Vrms or 1.4Vrms.
There are R&C networks for F compensation to give flatter response for all F ranges.
The Wien bridge PFB
network loads the amp with Zin = 2.112 x R where R is
the network R value. So if R max = 52k, R min = 2k2, then load changes between
109k and 4.65k for each F range. The NFB network with 12 series lamps has each
lamp operating with 0.46Vrms each which raises cold R or 27r to about 55r.
So the output side of NFB network has R = 660r, and slightly variable if applied
voltage changes. The input side of NFB network has an adjustable R which is used
to trim Vo and THD. Once variable R is set, it becomes fixed, and is twice the
average lamp R value, so 1,320r, so lamps plus fixed R = about 1,960r total.
When in parallel to PFB at highest F the total RL = 1.3k approx.
suitable lamps are the type "327" rated for 28V x 40mA, 1.1W, and has
= 60r and good for Wien bridge use.
should always choose an
easily replaceable lamp and make sure excessive voltage cannot occur in
During operation, minute Vo change causes minute change of lamp current and
lamp resistance. This alters the voltage fed back to input, ie, ß varies.
ß is NOT constant.
If Vo rises, ß increases, open loop gain is reduced. If Vo reduces, ß is reduced
and OLG increases. The PFB ß remains constant, 0.3333 for the output frequency.
For F above or below Fo, the PFB ß is always lower than 0.3333 and lower than
NFB ß so the circuit only oscillates at Fo ( providing there are no parasitic F. )
With both PFB and NFB applied the oscillations remain at a point of equilibrium,
and Vo is unable to increase or decrease.
The Vac difference between applied PFB and NFB drives the amp to produce Vo.
At frequencies above or below the output F there is less PFB applied,
and NFB acts to reduce THD and noise.
The open loop gain of amp
= x 355 (about
+50dB) and from V1+V2 12AU7
differential pair with
cathode CCS followed by 6AC7 pentode.
input pair has the V1 PFB triode with anode load of 560r and its
= 100Vdc. The input C for PFB is low, like a cathode
follower. The 560r may
be omitted, but I recall it is for HF stability. Differential gain of the 12AU7 is
8.0 and 6CA7 gain with unbypassed Rk = 42, hence total gain 335. The bypassing
of the Rk of 6AC7
would much increase gain and also increase problems with stability.
As it is, the non
bypassed Rk much reduces the THD of 6CA7 locally, leaving less
mess for the NFB loop
to clean up.
The open loop gain
is reduced to 3.0 with NFB and so gain at mid band at say
is reduced by NFB by a factor of 3 / 335 = 0.00895, or by about -
THD is at its minimum, and less than 0.5% and you cannot see it
using a CRO. Tests revealed it is mostly 2H.
THD in such
oscillators does not follow all normal rules for THD in amps with
because where you have say 3H being fed back from Vo it is phase shifted
and its difficult to fully calculate results. My calculations suggest THD is reduced
by a factor 0.083, or by approximately -22dB, and thank you Mr NFB,
accept your gift!
practice, the measured THD is seldom equal to the theoretical
there is some THD generated by lamps and by limiting diodes just beginning to turn on.
If one wants
extremely low THD with Wien bridge then it is most achievable where
the oscillator is dedicated to 1 F as in the case of one WB oscillator I made
with an op-amp giving 0.004% THD at 1kHz.
See my page at
Now for a much better WB oscillator using tubes for sine waves
and SS for square waves....
Picture 2. Front of tubed WB oscillator.
This was nearly completed 2013. Dials and finish were completed later.
The paint finish is a bit rough but what is inside the box is important.
Notice there are two F dials. Left dial is for AM radio tuning gangs, and right dial
is for an Alps pot for the lower F.
The dials and labels were made using a PC. A cardboard template with all F markings is
scanned to make a .gif image in MS paint and in black and white only.
The scan was tidied up in MS Paint. The tidied dials were printed and altered until
I got the sizes about right after printing.
Because the two frequency dials are circular, they can be slightly bigger or smaller
than originally intended and the pointers will still point to the same F numbers.
Three printed paper
copies were made, then overlaid in exact position and soaked in
furniture vanish which also acts as a glue to metal case front. Overlaid copies
gives depth to the printing and looks much better than using just one varnished copy
with allows old numbers and metal defects to be seen through the opaque varnished
Sheet 1. The more capable Wien Bridge oscillator in Picture 1 at top of page......
shows the result after many experiments with different circuits
to get maximum possible F = 2.2MHz.
trials with triode input stage LTPs with 12AU7 and 6DJ8,
the use of
6BX6 / EF80 were found to be the best for both gain stages, despite
the complexity of cathode biasing and screen bypassing.
frequency open loop gain of V1 & V2 at 1kHz = 22 x 30
The gain of the white follower is about 0.75x with all loads
used, so total open loop gain = 495.
reduces open loop gain to 3.0 so the amount of NFB = 44dB
loop gain below 30Hz and above 10kHz is reduced with the gain
and bypassing of V1& V2 470r cathode
resistors with appropriate R&C to give LF stability.
-24.7Vdc rail enables the use of high value Rk which helps to
keep the anode Vdc
very stable, so direct coupling of V2 anode to
White follower V3 grid is possible to avoid
the phase shift.
It is not
until anyone tries to build something like this that it becomes
difficult to see how a
simulation program could predict the exact
R&C values better than I have with an old brain.
But I did draw Nyquist graphs of open
gain and trials of R&C values and I spent much
time testing before I was satisfied the circuit would not oscillate, yet give me a passable
square wave at 2MHz.
I could get the initial circuit to be stable after initial warm up, yet half an hour later when
using the LF range I'd get bursts of HF parasitic F above 12MHz. I had to add an extra
wafer S1c to rotary switch, and add C4a 2n7 to reduce HF gain while using LF range.
I had a number of identical 3 gang AM radio tuning caps and I was able to mount them
all together using 4mm plastic and plywood and rotating them with dial cord around
4 x 65mm dia 20mm thick plywood discs glued to cap shafts. A dab of glue hear and there
made the dial cord idea work well to move all C gangs together.
The alternative would be fine chain wheels and tensioned chain but I found nothing
available and able to be adapted.
The marine ply discs were easy to make using a plumber's hole cutter blade in a drill press.
Each C gang gives C range approximately 13pF to 400pF. The effective C range of each
used in the Wien network for PFB is approximately 180pF to 2,580pF which includes
This meant having low R value of 253r for the 200kHz to 2MHz
range. The minimum C
of about 200pF including stray circuit C and the low
253r gave the F extension to 2MHz
and with stability. All other attempts with
and less C failed to give good
circuit was difficult to stabilize at both LF and HF to get the full bandwidth 1Hz to 2MHz.
The S1 range switch I have used was an old well
made switch from 1950s with 3 poles
and 9 positions and I have used 6 positions after
adjusting metal work to only
give the 6 positions. S1a and S1b change the range F, and S1c
just switches in the 2n7
to 0V, via link X to X.
So why use so many ganged tuning caps?????
To get up to 2.2MHz the minimum C needs to be above about 180pF or you get all sorts
of HF or LF instability. If C min = 180pF, then R must be theoretically = 401r. But you
can see I have 253r. So there must be considerable unknown stray C in the circuit,
maybe 105pF. If C is any less than I have it, the circuit won't work very well, and for higher
F you would need to think about much lower R and still keep C highish and use higher gm
devices such as RF signal mosfets etc. I could not find any schematics online for Wien
bridge oscillators able to go to say 22MHz.
If I wanted a decade of F change per range, and C min was 285pF including stray C, then max
C would have to be 2,850pF. To get this C variation I needed lots of variable caps.
Then to get down to 1Hz, the R size = 55.8Meg ohms. Well, that's quite an impossible R value
because the minute grid current in V1 ruins the function and you find you just cannot get down to
1Hz with tuning C unless you had R not more than about 5Meg. I settled for lowest F with
tuning C = 20Hz which allowed R = 2.8Meg. I wanted to have fully variable F and a pot for
lowest F range with fixed C was more sensible, and I could get from 20Hz down to 1Hz.
I could have had 24 values of switched C or R but I would not have "tunability" in operation.
THD is fairly constant at < 0.4% from 100Hz to 5kHz. It would
be possible to have far less
THD but then there are problems of trace bounce and stability,
so the circuit has TWO
networks to manage the problems. There are 12V zener diodes plus
1N4007 to limit the
positive and negative FB from the output of V3&4. Then I
have two strings of 14 x 1N914
in series, facing in both directions so that once the Vo rises
too much, the diodes conduct
and apply MORE NFB to the 4 series lamps and NFB applied to V1
immediately without waiting for lamps to warm and increase their
happens too slowly, and leads to trace bounce.
Without diode clamping, the circuit tended
to oscillate at both LF and HF when Vdc
conditions changed slightly and briefly at range
switching, and at high end of the 20Hz
to 200Hz F range where grid biasing resistance for V1 is 2M8 in
the PFB network.
The phenomena may be called hysteresis, (( harder to deal with
than a hysterical wife )),
and the phenomena affects all amplifiers with high open
loop gain, several stages of
tubes which draw grid current at mild overload, lots of RC
couplings, large amounts
of PFB and NFB loop and a NFB loop. I was amazed that the damn thing worked at all.
It is very easy to make a Wien bridge
oscillator with range from 20Hz to 200kHz.
The difficulties occur when you aim for 1Hz to 2MHz.
The diode clamps with 14 x IN914 in series exploit the gradual
at forward voltage of about 0.4Vdc when current is less than
2mA, and each diode
becomes a low resistance at 0.72Vdc. With 14, the V
range for threshold turn on is from about
6Vpk to 9.8Vpk. So the diodes form a non linear
controlled resistance where an
increasing applied voltage causes a logarithmically increasing
current, and lowering of
effective resistance. This action rapidly increases the NFB
voltage delivered to V cathode
before the lamps have heated up to increase their resistance. The lamps have
slow time constant and diodes have an instant time
constant, so the diodes give an
immediate increase of NFB to prevent the instant rise of PFB
from causing excessive
Vo to occur. The circuit settles quickly between range
switching, and once settled,
diodes have only a slight effect while lamps take over the major
part of the NFB regulation
of the Vo.
At Vo level of 7.5Vrms, some 3H begins to be produced by the
diodes, but it is better to
have that to enjoy the easy stable operation as a
result and without lengthy settling times
and jittery Vo amplitude. I found that if the Vo level was
to say 6.4Vrms by adjusting
VR2, the THD < 0.1%.
look at the above Sheet 1 schematic and be repelled by the
and massive amounts of RC "damping" networks, but if you
analyze each stage and even
draw the open loop response then it all makes sense. If you want
such wide bandwidth,
reasonably low THD, but with complete LF and HF stability, then
complexity is the price.
The open loop phase shift never exceeds
about 100 degrees. With gain reduced by
gain shelving networks the amount of NFB applied at 2Mhz and at
0.5Hz is very much
lower than at 1kHz, so you get stability. The tubes are
inherently linear at their normal
signal voltage levels. The increase of THD at
extreme HF and LF is difficult to see on
the oscilloscope and of no concern. The F response between
within +/- 0.5dB.
The PFB and NFB networks create the load on the White follower output buffer stage.
With variable C and fixed R in PFB network the minimum PFB load = 2.11 x 253r = 534r.
Lower F ranges give higher load ohms with less THD and gain reduction.
NFB network load is 180r in 4 series lamps plus 360r in VR2 and surrounding R giving
total load NFB load = 540r.
Therefore both PFB and NFB networks give lowest total load = 270r. The maximum
Vo is determined by the minimum loading and idle current of output buffer.
Vo max = 7.5Vrms so if load = 270r, then Iac = 28mArms = +/- 40mApk, and idle
current in White buffer = 56mAdc so there is adequate Ia for class A in output tubes.
EL86 in triode needed to have Iadc at idle
of 56 mA and Ea = 125Vdc, with
Pda = 7 Watts each, and less than their 12W rating. The EL86 works
EL84 because EL86 has 1/2 the Ra of EL84, hence is less
affected by the change
of total RL across each F range, caused by change in PFB network
The gain of
the follower begins to decline slightly in the top F range but the NFB boosts
signal to compensate.
V2 has to make a maximum of about 11.5Vrms, well within its SE
cut off or grid current.
3, 4 give details for what follows the sine wave oscillator.
is a switch for sine wave or square wave, a solid state discrete
Schmitt trigger square wave generator with its own amp for +/- 10.1Vpk.
There is an emitter follower buffer to
power a low resistance output attenuator
switch for 6 levels of
-10dB steps, and a 300r linear output pot to output
Sheet 4 gives PSU details including voltage regulators.
Sheet 2. Output buffer and
Sheet 2 has npn and pnp bjts in a pair of Darlington connected complementary
emitter followers to make a buffer stage after sine wave or square wave sources
to drive the low resistance output attenuators without losing any bandwidth or
stage is needed to avoid loading the sine wave or square wave
circuits with low R attenuator networks. and to prevent connections with outside
devices from having any effect on the signal production of the
output load for the buffer could be 47 ohms if short circuit
and 0V occurs with switched attenuator in position
1 and pot VR5 is turned up for
maximum Vo. A square wave of +/-10V peak from buffer will create
+/- 212mA peak, so each pnp and pnp output bjt must have Idc at
idle of about
120mAdc to allow *class A* power even when Vo is
shorted to 0V.
With +/-10Vpk square wave output the 100mA fuse will blow.
Pd in R14 47r is 2.1Watts and rating is 0.12W so R will blow if fuse does not,
For all other
switch positions buffer load is higher ohms and signal all class A
and THD very low.
To avoid damage by accidental connection of unit output to damaging voltages,
ie anode B+ supply, or mains active input, the network with R14 47r 0.125W,
100mA fuse and diodes to +/-15V act before allowing damaging current into
all signal sources above 100kHz should have low impedance of 50 ohms
avoid effects of cable or device input C or L. But because I have gone up
to only 2.2MHz and not to 10MHz, then a 300r output pot is OK.
change to the square wave occurs at F above 200kHz, regardless
of level setting.
of the bjt buffer is from DC to 12MHz at least. A CRO shows no
change to the shape of 2MHz square wave.
The rise and fall of square waves at 2MHz looks like the highest harmonic is about
5MHz. This limited H content is due to simple Schmitt circuit which was used instead
of other more complex or delicate IC parts which would not produce the
+/- 10Vpk I have here. There is little use for square waves over 100kHz to what is
here is just fine, and at 5kHz the square wave is better than nearly everything else
I have ever seen.
Sheet 2 has a table on how to calculate the resistance values for
switched attenuator network. This network is very useful for getting an easily
adjustable wide range of output voltages.
preamp can be easily be supplied with a 5mV of input signal.
All good phono amps for MM or MC carts should survive
having 5mV at 20Hz,
50mV at 1kHz, and 0.5V at 20kHz.
available at maximum level setting, one may test the F response
OPT and most low Z inputs of say 600r, knowing the Rout
= 50r, and quite low enough.
Sheet 3. Square waves with
Sheet 3 shows a Schmitt trigger sine-wave-to-square-wave converter
by a small signal discrete bjt op-amp
which I developed especially for this project.
Schmitt trigger circuits using tubes consist of a single 12AT7
half used in a similar way to Q6 and Q7 in
Fig 4 above. But they struggle to make a
good square wave without reducing rise times and "rounding of corners" of the
ie, HF content is attenuated. So while a 10kHz square wave may have F up to
the 100kHz square wave will have very poor
shape so I did not try to use them.
Perhaps a pair of EL86 or EL84
in pentode mode might work better than a 12AT7.
YOU to try, and I wish you good luck, you will need it.
Here is a
result of spectral analysis performed on a 1kHz square wave...
for a perfect square wave the odd number harmonics would have to
to an infinite frequency well beyond the ability of
home brewed test gear to achieve.
But a good
looking square wave will have 21 times the lowest frequency
is called the fundamental frequency, aka H1, or Fo.
With this in mind, for testing HF
stability of amplifiers, use of a 10kHz square wave should tell
us if there is an unwanted
peak in the F response below 210kHz, something
very common in all too many awfully
engineered amps. But seldom will 21 times H1 exist in
any square wave, and at 2MHz,
the attenuation of the 10MHz content and beyond may be below
-35dB and all other
higher H buried in noise.
So to get
anywhere near the ideal spectral analysis shown above, we need
switching of devices used to produce the waves. I tried all
sorts of Schmitt triggers
with bjts based on what could be found in books and online.
Always the performance
satisfy above about 350kHz, and failed to function
properly by 2MHz.
Googling for an easy way to generate a 10MHz square
waves from a sine wave
using a few simple devices gave no solutions.
But there were
very complex schematics
with many ICs which like others
online probably failed to
include all associated circuitry to give certain results.
I tried a 4093 C-mos chip, and all ran right out of puff by about 300kHz, and
none could have the necessary trim pots adjusted to give a symmetrical square
wave for all F between 1Hz and 2MHz. I
concluded mosfets would be best
because they don't conduct current at their bases like
PN100 or PN200 bjts
especially at HF that they become so mis-biased
by 300kHz the circuit becomes
and there is little wave symmetry and H content is poor, and Vo level
sags. Sheet 3 shows
exactly what I found to be
best using a pair of 2SK363 driven at the input by a PN100 in
emitter follower mode.
fixed bias of +2.8Vdc at the second mosfet and there are TWO
one for input Vdc bias and the other for the level
of input. The circuit works well
right up to 2MHz, without difficult R&C compensation
networks or super critical
of the trim pots. The 2MHz wave isn't all that good, and it
a 10V peak sine wave which has been clipped down to
4V peak with slight corner
rounding. But at 100kHz, the square wave appears to have H
extending to 2MHz, or 20 times the Fo, so good enough.
circuit can make
only about +/-2.8V peak. Increasing Vo with a following amp might reduce H content.
But the amp has BW = 0.3Hz to 12MHz, and good enough. There is a network
R11 + C8 between trigger and amp input to
stop parasitic H above 20MHz
between mosfets and
Whatever networks you see used have all been
trialed for best component value.
But this simple amp with 6 bjts works better than nearly all op-amp
can buy at Jaycar,
Dick Smith, or The Radio Shack etc.
high Vo of +/-10V peak, there is an opportunity to use an
clipping circuit with a low value R of say 470r and
a pair of 1N914 diodes in
parallel in opposing direction to give +/- 0.7V peak. Use of 4
make +/- 1.4V peak. This may then produce a
square wave of sufficient amplitude
to make many power amps clip, and including much extended
I have not fully explored possibilities, but fast diodes UF4004, UF4007,
Schottky 1N5819, are easily available.
Sheet 4. Power Supply.
Sheet 4 shows a generic PSU which need not be exactly the same as any
you might build but you just need make up the voltages and currents as
shown and with the regulation shown with equal functionality.
One should be careful with discrete bjts used for HV regulation because
all too easily the 3 bjts use above can all instantly fuse
to a sullen short
circuit needing 2 hours of work to replace them. Hence the protection
you see, and which you don't always see on schematics
posted online by
idiots who never consider that stuff fails after shit happens. YOU need
rails are essential to stop very LF noise appearing in the
Such noise is mostly under 10Hz, but its presence
with tests on
amplifiers, so it cannot be tolerated. The LF rail noise is created
unavoidable changing mains voltage levels of other mains users switching on
devices continually. Solid
state regulators are so much easier to make than
The +266V regulator works very well. There
good insulation between the Q1 BU208 heatsink lest a
tiny little arc occur
which will cause instant bjt death.
If anyone want to make a discrete part op-amp for a general
small signals up to 10MHz, they may consider
Fig 2. Small signal amp,
Fig 2 is basically the same amp as in Sheet 3.
But here it is as I developed it and there are precautions about
It should be used with loads above 600r. It may not like
R19 100r increases its Rout, but reduces effects of C loads.
of C6 & R12 will give best HF extension depending on the
load to be driven.
You may have to add a 220r with 33pF LPF at input to reduce
oscillation when used with a preceding amp. But it is food for
with your soldering.
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