SINGLE ENDED OUTPUT BEAM TETRODES.

This page
explains how to set up
single ended output beam tetrodes in tube amplifiers

using the 6550 and KT88 working in pure class A beam tetrode
mode or
with 43%
UL

screen taps, or with local cathode feedback windings which are a
portion of the OPT

primary winding.

This page
has the following
content :-

Operation of the beam tetrode with cut-away sketch
by RCA.

Fig1. Graph of Ra curves
for GE
6550A beam tetrode with
screen supply = +350V.

Fig 2. Graph of Ra
curves for
GE 6550A beam tetrode
with screen voltage = +350V

plus 3 load lines and calculated results for gain, power output
and
second harmonic

distortion.

Fig 3. Graph of Ra
curves for
GE6550A beam tetrode but
with screen supply = +200V.

Fig 4. Graph of Ra
curves for
GE6550A with one load
line for 3.5k and calculated

gain, power and 2H.

Explanation of the effects of NFB application for 6550 beam
tetrodes.

Formulas for NFB and output resistance.

Fig 5. Graph for power
output
for single 6550 beam tetrode vs anode
load value.

Choosing the OPT ratio.

Fig 6. Graph of 6550 Ra
curves
for 6550 in UL mode with
43% screen taps.

Calculated gain, power output and 2H for 4 different load
values.

Data for Ra, µ and gm for UL and comment on the effects of
NFB

and distortion outcomes.

-------------------------------------------------------------------------------------------------

This tube has four electrodes which are cathode, k, control grid, g1,

screen grid, g2, Anode, a, and two beam forming plates near the anode,

with each connected to the cathode.

but in fact electron flow occurs on both sides of the cathode.

The central cathode consists of a flat metal tube and inside this tube is a length

of folded filament wire coated with a special inert insulating metal oxide to

prevent current flow from heater to cathode. The filament has 3.5 ohms

resistance when hot with an applied 6.3Volts at 1.8Amps needed to raise

the temperature of the cathode to about 900C.

The small
orange glow seen in
tubes is the hot central cathode. All
other parts are

designed to run at a much lower temperature without any glow of
their
own.

The cathode, k, is often
nickel and coated with special metal oxides which promote

the emission of a cloud of electrons once
the tube is warmed up. This emission is

only possible if there is a very pure vacuum inside
the sealed glass tube structure.

The grid,
g1, is made of special
wire wound in a slightly flattened helix and wires

are spot welded to two support rods. All the grid support rods,
anode
and cathode are

supported in discs of mica sheeting which is inert and which
rests
against the glass to

locate all
electrodes at their correct positions within small tolerances.
The
pitch and

size of the helical coil and and its
distance from the cathode determine the tube

behaviour.

Grid 1 is usually set up with a bias dc voltage
about -50Vdc below the cathode

potential. By varying the grid 1 voltage + and - with an ac
signal, the
flow of electrons

from cathode to anode is controlled fairly linearly. The grid 1
draws
no current
because

its negative charge repels negatively charged electrons.

There is
a second grid of larger
size, the screen grid, g2, placed outside the grid g1

and with its
helical coil wires aligned to the g1 wires to allow about 90% of
electrons

to
flow past and be absorbed by the anode. The screen g2 usually
has a
potential

at +300Vdc above the cathode and
it attracts 10% of anode current flow and the

heat generated is about 1/15 of the heat generated at the
anode.

The TWO
beam forming plates are
connected internally and permanently to the

cathode. Their function is to repel electrons so that the
electrons
which are attracted

to the
screen and anode are focused into two beams on each side of the
cathode

The presence of the beams of
concentrated electrons suppresses the effect of electrons

bouncing off the anode and returning to the next
best positive thing, the screen, which

is called secondary emission. The beam forming plates were
invented to
counter the

need to pay royalties to earlier European inventors of the
pentode
which has a third

grid,
g3, between screen and anode which functions similarly to
supress
secondary

emissions which otherwise
spoil the function of pure tetrodes.

The plate, or anode as it should be called is a sheet metal box
with
open ends and

it absorbs about 90% of the electrons emitted from the cathode
with the
balance of

10% being
absorbed by the screen. The anode has open ends to allow heat to
radiate

from the cathode more
easily up and downwards. The anode connects to the load

being driven by changes in Va and Ia.
The anode is heated by the flow of electrons

and its heating is known as its plate dissipation, Pda,
easily calculated as Ia x Ea.

The
silver coating on the inside
of the glass is called gettering, a very thin layer of

special metal
which absorbs stray gas molecules to form compounds which are
then

locked to
the glass rather than re-cycling around the tube and destroying
its
operation.

The
advantages of the beam
tetrode and power pentode allow for about twice the

audio or RF output power to be generated when compared to
triodes of
the same

size and
cost of production and same anode dissipation. The beam power
tube or

power pentode can be used
with its screen grid connected to the anode to make the

tube operate as a triode. In the case of the
6550, the maximum anode dissipation

rating is 42 watts, and the theoretical maximum amount of audio
power
from a single

tube in class A1 = 18 watts in beam power mode, but only about
10 watts
in class A1

triode. In practice, nobody should ever have a single 6550 set
up with
Pda = 42 watts

at idle because the
tube would run too hot and have a short life. Pda plus Pdg2
should

never exceed 35 Watts at idle, and
in fact 28Watts is optimal for class A1, and perhaps

22W for class AB. With Pda = 25W, max power in class A
SE beam tetrode = 11 watts,

and in triode about 7 watts. It is possible to extract more
power in
triode by operating

in class A2, and get perhaps 10 watts but I feel there are not
many
benefits because

of
the higher THD and cost of extra circuit complexity.

Not many
people will ever have
access to a supply of NOS GE6550A or NOS KT88

which are very similar. Most old tubes made in the US before US
factories or UK

factories closed have now been worn out. So tube supplies will
be from
Russia,

China or eastern
Europe.

The New
Sensor Corp sold me a
large batch of EH 6550 tubes in Feb 2002.

My previous update of this website was in March 2006, and I have
not
had any

reports of any early
failures tubes in that purchase. It is now April 2011, and I

still have no reports of early failures.

The
working characteristics of
these recently made Russian tubes appear to be

equal to NOS samples of the same tube type such as the renowned
GE6550A,

and can be generally
be used as replacements in all amps requiring 6550 or KT88.

I also recently measured Sovtek KT88,
which gave identical test results to the

EH6550. Sovtek or EH 6550 or EH KT88 from Russia all
appear to me at least

to have exactly the same internal physical structure and
electronic
beam tetrode

data
parameters of Ra = 19,000 ohms, Gm = 10mA/V, and µ = 190
at Ea =

+400V, Ia = 90mA, and Eg2 =
+350V.

But
usually 6550 or KT88 are set
up with Ea = 450V, Eg2 = +350V, Ia = 51mA,

Ig2 = 3mA. Then Ra = 32,000 ohms, Gm = 5.5mA/V, µ = 176.
This is
very

different to data quoted for the higher Ia condition which
causes Ra to
be lower

and Gm to be higher.

Anode
curves for GE 6550A are
similar to EH6550.....

Fig 1.

These Fig 1 curves are
for
GE6550A but typical for any 6550.

If we
look along the Ia = 100mA
line we see that for a change in grid voltage

of 1.5V there is an Ea change = 350V, so µ = 350 / 1.5 =
233.

If we look along the Eg = -17V line we see that for 350V of Ea
change
there is

an Ia change of 18mA, so Ra = 350 / 0.018 = 19,444 ohms.

Therefore gm = 12mA/V.

At Eg =
-10V, the Ra = 10k and
µ = 100 so gm = 10mA/V.

The anode
curves are more
crowded together vertically at the bottom of the

graph indicating a large change in Ra, µ and gm between Ia
= 50mA
and Ia

= 300 mA.

Here are
some load lines and
analysis for loads of 1.75k, 3.5k, 7k....

Fig 2.

Here we
have the 6550 loaded
with 1.75k, 3.5k and 7k. Some of you may

not know what the heck a load line is.

The load
lines in the above Fig
2 are straight lines drawn across the data sheet

"anode characteristic curves" and such straight lines represent
load
ohm values.

For example,
consider load line for RL = 1.75k. It passes through the voltage

axis at +512Vdc, and also through the Ia
axis at 293mA. The "slope" of this line

represents a load resistance = Voltage / Current = 512V /
0.293A = 1,747 ohms,

or close enough to 1.75k.

The idle
point of the 6550 is at
Q where Vdc between anode and cathode

= +350Vdc. This a - k voltage is known as Ea. The idle Iadc for
the
6550 is

at 90mA.

This means Pda
in this example = Voltage x Current = 350 x 0.09 = 31.5 Watts.

The load
line for a single ended
tube MUST pass through the Q point. It will

intersect the curves for Ra at various values of negative grid
voltage.
You may

wonder how do
such curves come to be, but its simple, really.

To
determine 6550 beam tetrode
anode curves, the anode is connected to a

low impedance signal voltage supply able to swing to perhaps +
and -
600Vpeak.

Such a signal
may come from a 50Hz 424Vrms transformer winding with one

and grounded and the other to the
anode. The cathode is grounded, and grid has

a variable negative supply connected to give a
range of between 0V and - 50Vdc.

There is a 50 ohm resistance connected between the 424Vrms
winding and anode

to allow a second small transformer to transfer the Vac across
the 50r
to one
trace

of an oscilloscope in X-Y mode so that the amount of current
flow in
the 50r anode

current
sensing R is related to to the applied and varying anode
voltage. A
sample

of this voltage is applied
second trace of the CRO. The curves may be seen as

those above for various values of grid bias
voltage. Machines to automatically

draw such curves were developed many years ago for use in
tube development

laboratories. So the curves are a way of depicting the
dynamically
changing anode

resistance, Ra, because the curves are simply describing a
function of
Ohm's Law,

where
I = V / R.

power and second harmonic distortion.

The grid -Vdc bias at Q1 = -18.5Vdc. If we look at the 1.75k load line, we can see

where it intersects the Ra slope at Ia = 270mA, and Ea = +40Vdc. The high slope

line close to Ia axis is the limit of linear anode voltage movement. Beam tetrodes

( and pentodes ) are quite queer because the Ra is initially very low and only 225

ohms for where Ea < +50Vdc, and for higher Ea voltages the Ra suddenly becomes

thousands of ohms. But the queerness is acceptable and it allows a good wide useful

Ea voltage swing.

At Ea = +40Vdc and Ia = 270mA, grid voltage at this point = -3Vdc. We have

changed grid voltage by +15.5Vdc. Load voltage has changed by -310Vpk. If we

change grid voltage -15.5Vdc to -33.5Vdc by applying a linear undistorted grid input

signal then we may read from the load line where the grid voltage = -33.5Vdc, and

it is where Ea = +495Vdc. Load voltage has changed by +145Vdc.

If the tube was perfect, and it surely is not, the changes in Ea and Ia would be equal

in both positive and negative directions. From the above we may calculate 2H distortion

as 2H % = 100 x 0.5 x difference of V swings / sum of V swings = 50 x 165 / 455 = 18%.

Output Load Vrms = Vpk-pk / 2.82 = 455 / 2.82 = 161Vrms and

PO = V squared / RL = 14.8Watts.

Class A efficiency % = 100 x PO / Pda = 100 x 14.8 / 31.5 = 47%.

This is a very poor THD outcome, but the 1.75k is a POOR LOAD MATCH, and

the lesson here is that we should try to get a better load match if we want better

sounding hi-fi!

So, with the Fig 2 Q point, we get......

1.75k :- Vg = 11Vrms, Va = 156Vrms, gain = 14.2, power = 13.9 watts, 2H = 19%.

3.5k :- Vg = 6.7Vrms, Va = 189Vrms, gain = 28.2, power = 11.6 watts, 2H = 12%.

7.0k :- Vg = 4.7Vrms, Va = 216Vrms, gain = 45.9, power = 6.7 watts, 2H = 3.7%.

The
"best" RL = 3.5k, and at 1
watt, at Va = 60Vrms, and THD is about 3%.

From
these rather poor figures
we can see that 2H is dreadful at low RL but reduces

as RL rises. In fact 2H will continue to reduce to 0% where
there is a
null in 2H

production and then the 2H will begin to increase as RL
increases. But
the relative

phase of 2H will be
opposite to phase at low RL. The load lines will show that if

you care to interpret them correctly.
The figures for 2H don't include most of the

other odd and even number harmonics which are significant
and numerous. The

awful linearity of beam tetrodes and pentodes may be tamed with
NFB,
but
first

we need to just consider loadlines.

Here are
curves for GE6550 with
screen voltage a lot lower = 200V :-

Fig 3.

Sometimes
the beam or pentode
tubes produce better curves so lower distortion

with screen voltage lower than anode voltage.

Here is a 3.5k load plotted on the above curves :-

Fig 4.

With a
lower screen voltage =
+200Vdc, The operating point can be Ea = 350V,

Ia = 90mA for Pda = 31.5 watts as in the case where Eg2 =
350V.
The load

is
3.5k.

The outcome at maximum power with no NFB :-

Vg
= 6.7Vrms, Va =
185Vrms, gain = 27.6, power = 9.8watts, 2H = 8.3%.

At one watt, THD = 3%

If there was 20dB of global NFB we could expect THD at 9.8 watts
to
become

about 1% and 0.3% at 1 watt. Maximum power would be boosted to
about 13

Watts
because the anode V swing would be increased to about 600V pk to
pk

from the 523V shown
above. There is an increase in power output which is

proportional to the
increase in V swing squared.

Use of a
beam tetrode shows that
the load that gives a healthy power output

also produces high THD even at low levels, because THD is
approximately

proportional to
output voltage. Where you have 9.8 Watts of output, Vo =
185Vrms,

and where you have 0.98
Watts Vo = 58Vrms. The change in THD will be from

8.3% to 2.62%.

I would
never use any beam
tetrode pentode in single ended mode unless I

could have ultralinear taps or cathode feedback windings on the
output
transformer,

or I just
use plain triode connection.

Let us
suppose the OPT ratio
chosen was 5k : 6 ohms, or 833 : 1, so that if there

is a dip
in speaker Z below the nominal 6 ohms to say 4.3 ohms, the load
seen by

the anode = 3.5k as shown above. The high Ra of the tetrode =
approximately 17k

ohms at the
working point Q. The Ra of the tube measured at the secondary
=

17,000 / 833 = 20.4 ohms =
output resistance without NFB. The damping factor

= RL / Ra = 6 ohms / 20.4 ohms = 0.29, and
atrocious, when what we want is

DF = 10 if possible.

Let us
now explore the effect of
20 dB of global NFB on an SE tetrode amp.

Where 20dB of global NFB is used, the gain without NFB is
reduced by a
factor

= 0.1.

Consider
the SE 6550 in tetrode
in the above load line sample with 4.3 ohms

load on the OPT secondary. we have 6.7Vrms grid signal to make
9.8
watts into

4.3 ohms
= 6.5vrms. Say have a 12AX7 used to drive the 6550 and it has a
gain

of say 67.
Therefore 0.1Vrms is needed at the 12AX7 grid to produce 6.5
Vrms

at the load.

Therefore
gain without NFB,
called Open Loop Gain, OLG, = 6.5 / 0.1 = 65.

Suppose we reduce the OLG by applying 0.9Vrms of NFB to the
12AX7
cathode

via a
low resistance divider network from the OPT secondary speaker
connection,

using a typical
divider of R1 = 293 ohms plus R2 = 47ohms.

This will
reduce the apparent
over all gain with FB applied and is known as Closed

Loop Gain, CLG. CLG will become 6.5, 1/10 of the OLG of 65.

But let
us use RDH4 terms, and
OLG = gain A without NFB, and CLG = A' with

NFB applied.

ß,
the fraction of the
output fed back = R2 / ( R1 + R2 ) = 47 / 340 = 0.138.

The formula
for working out A' where we know A and ß is :-

A'
=
A
/
( 1 + [ A x ß ] ).

Distortion
with
FB,
Dn'
= Dn /(
1 + [ A x ß ] ),

where Dn = distortion without FB.

A' here =
65 / ( 1 + [ 65 x
0.138 ] ) = 6.5.

The
amount of NFB in dB due to
FB is 20 x log (A / A') =

20 x log (65/6.5) = 20dB

We can
say THD of 8.3% would
reduce to 0.83%

But if
there was no load on the
tetrode output tube, open loop gain,

A = ( gain of 12AX7 x µ of 6550 ) / OPT turn ratio = 67 x
190 /
28.9 = 440.

If open loop gain was 440, and ß remained at 0.138, A',
closed
loop gain =

440 / ( 1 + [ 440 x 0.138 ] ) = 7.12. The amount of gain
reduction is
much

greater where A is much higher.
The change in A' of between 6.5 and 7.12 is

not much, so tubes with varying gains will have much
more even and constant gains

when NFB is used. So NFB with two channels of SE amps will keep
the
balance

constant and give us better stereo imaging.

With a
load of 3.5k, the
feedback resulting from ß = 0.138 gives 20dB of gain

reduction,
but without any output load at all there is a gain reduction 440
to
7.12

= 61
times, = 36 dB of effectively applied NFB. This amount of NFB
**will**

make the amp become unstable at both ends
of the audio spectrum and is a reason

why steps must be taken with beam tetrodes and pentodes to
use zobel networks to

reduce gain and phase shift in the open loop character below
20Hz and
above
20kHz.

When 20dB
of global NFB is
applied with the 3.5k RL the open loop distortions

are also reduced by approximately 20dB. 2H, which is the main
harmonic
product

will be
indeed reduced about 9 times to 0.92% at about 9.5 Watts and
0.3% at 1
Watt.

However, since the 2H
is fed back there is a slight production of 3H by means of

intermodulation and there is greater
spectral complexity of the harmonic content

after FB is applied then before, even though the level is much
reduced.

The
effect of intermodulation
where only 10dB of NFB is applied around a tetrode

amp is all the more with an increase in 3H that could make the
sound
worse.

So when
applying NFB, use enough where distortion is high to begin with
as

is the case with a tetrode
amplifier.

The
calculation of output
resistance is more difficult to calculate because we are

need to take
into account the µ of the output tube.

To reference our calculations to
the secondary connection,

µ of the output tube at the OPT secondary = µ / OPT
turn
ratio.

In
this case it is 190 / 28.9 = 6.57.

The output resistance of a tube amp can also be calculated if
you know

1, voltage gains of the input and driver stages,

2, µ of the output tubes,

3, turn ratio of primary to secondary which is the unloaded
voltage
ration between

P and S windings,

4, the anode resistance, Ra of one output tube at the Q point.

5, winding wire resistance of the primary and secondary windings
of the
OPT.

Rout of the SE amp with FB
applied =
Ra
+ Rw total

ZR
x
(
1
+ [ A" x {µ/TR} x ß ] )

Where Ra
is for the one output
tube,

Rw total is the sum of OPT primary winding wire resistance and
ZR x
secondary

winding wire resistance. Allow Rw total to be 10% of the rated
primary
load if the

Rw measured
is unknown. TR is the turn ratio of the OPT, or unloaded P to S

signal voltage
ratio at 1kHz, or square root of the exact known ZR. ZR is the

output transformer impedance ratio which is
the turn ratio squared, A" is the

gain of the stages preceeding the output tube/s, ie, V at
output tube grid / Vg-k

of the input tube, µ is the amplification factor of an
output
tube/s,

ß is the fraction of OPT secondary voltage fed back to be
"in
series"

with the input voltage to V1.

For
example in this case we have
:-

Ra-a = 17,000 ohms,

Rw = 10% of primary RL of 3.5k = 350 ohms,

TR = 28.86 : 1

ZR = 5,000 ohms : 6 ohms = 833 : 1,

A" = 67

µ = 190

ß = 50 / ( 50 + 600 ) = 0.138

Rout' = closed loop output resistance at the secondary output
terminals

833 x ( 1 + [ 67 x { 190 / 28.9 } x 0.138 ] )

This is a
good result and could
be compared to the result with the 6550

connected in triode where Ra = 900 ohms, µ = 7.33, and the
OPT
had the same ratio.

But for 20dB of applied NFB with 3.5k anode load, ß must
be
increased to 0.73

and with the same driver tube with its gain 67.....

Rout',
triode,
=

900
+
350
= 0.11 ohms, 1/4 of the tetrode case.

833 x ( 1 + [ 67 x
7.33/28.9 x
0.73 ] )

We
can
conclude that about 17dB would be enough global NFB for
the
tetrode

amp to get Rout = about 0.5 ohms. The triode amp would need only
8 dB
to

achieve
the same low Rout.

off the tube curves. When measured, the distortion isn't low unless a lot of NFB

is used, so the graph is a very rough guide to power output and valid only for the

Ea & Ia conditions.

Fig 5.

The power
levels shown are
without any NFB and at the anode with no OPT

losses, and could be approximately 20% more with about 20dB of
NFB, to
give

a net 10%
increase allowing for an OPT with 10% winding losses. With NFB
the

2H and other distortions are
reduced to less than 2% and voltage swings on the

load are less limited.

The OPT
for the above Ea and Ia
condition should have the lowest expected

speaker load matched to 3.5k, so if the speaker's nominal Z is 6
ohms,
allow

for a dip in Z
to say 4.3 ohms so the OPT impedance ration = 3.5k : 4.3 ohms

which is 5k to 6 ohms, ie, 833 : 1.
The Rout before NFB is about 21 ohms and

with 20 dB of global NFB it will be approximately
0.33 ohms as calculated above.

A trioded 6550 will also give Rout = 2 ohms without any NFB but
with 12 dB of

global NFB the Rout will be about 0.6 ohms.

Single Ended Ultralinear could be another option to use.

The data here is for a GE 6550A with 43% screen taps.

Fig 6.

where 280 is a constant number of ohms which is based on the Ra limiting

line slope for Ra below the knee of the Ra curve for Eg1 = 0V.

The other traditional formula for SE pentodes or UL or CFB connected

tubes is RL = 0.9 x Ea / Ia.

One may wonder about the constant of 280 ohms. But look at the beam

tetrode curves above and the slope of the Ra lines to the far left side between 0V

and 50V have a 300mA current change, so the slope of the all the beam tetrode

lines below 50V = 50 / 0.3 = 166 ohms but I like to allow a little more, especially

for UL, because the UL curves are not so clear, hence the above formula and Ra

for beam tetrodes and pentode and for UL as seen above or for CFB need only

be 280ohms for OPT design. The actual E minimum swing in the above UL

graph is somewhat vaguely indicated between 0V and Ea = 75V.

If I were
to start with an Ea =
400V and Ia = 80mA for Pda = 32 watts,

RL = ( 400 / 0.08 ) - 300 = 5,000 - 300 = 4,700 ohms.

If the operating idle point is 350V x 90mA as
in the
above graph, we

get RL = ( 350 / 0.09 ) - 300 = 3,588 ohms.

With 43%
taps, the SE tetrode
characteristics change to the following :-

Ra = 2.0k, µ = 15.4, gm = 7.7mA/V at Ea =
350V,
Ia = 90mA,

Pda = 31.5 watts.

The performance outcomes are :-

1.2k
:-
Vg = 17Vrms, Va = 92Vrms, gain = 5.4, power = 7.1 watts, 2H =
15%.

2.0k :- Vg = 17.7Vrms, Va = 131Vrms, gain =
7.4, power = 8.5 watts, 2H = 12%.

3.6k :- Vg = 17.7vrms, Va = 170Vrms, gain =
9.6, power = 8 watts, 2H = 8.3%,

( 3.6k load line not shown )

4.7k :- Vg = 17.7Vrms, Va = 191Vrms, gain = 10.7,
power = 7.8 watts, 2H = 5.5%.

11.7k :- Vg = 17.7Vrms, Va = 218Vrms, gain = 12.3,
power = 4.0 watts, 2H = 1.7%

The "best" RL = 4.7k, and at 1 watt, at Va = 60Vrms, and THD is
about
1.5%.

The use of global NFB increases the maximum power output up to
20%
without

more severe THD.

The
advantage with UL is that
although the 2H remains fairly high, there is much

less other harmonics to worry about and the tube behaves much
like
triode but

without the limitation of the Eg = 0V limiting the Ea minimum as
much.

The load
for maximum power is
about 3.6k, and so nominal speaker loads

should match about 5k.

Therefore
an OPT would have a
ratio of say 5k : 6 ohms which has an impedance

ratio = 833 : 1. This allows for any dip in speaker load
impedance to 4
ohms without

hugely increasing distortion. The UL Ra when measured at OPT
secondary
=

2,000 / 833 = 2.4 ohms. 12dB
of NFB will reduce this further to about 0.8 ohm

and nearer to a triode with the
same global NFB and OPT ratio.

If we
plotted a load of 4,700
ohms through the Quiescent point Ea = 400V

and Ia = 80 mA, we would get a swing of about 0.9 x Ea = 360V
peak
giving 13.7

Watts
maximum in theory. In practice we may get 13 watts when pushed
into low

THD by NFB
application. If you plot low loads less than 2k the phase of the
output

2H is like triode, and
2H diminishes to zero at some load above 2k then 2H increases

as load increases further but phase of
2H is opposite. The SEUL connected tube

has only slightly less 2H than triode or about 1%
when power output is 1/10 the

maximum.

The use
of distributed loading
in the OPT with 20% of the primary turns in a cathode

feedback winding and with the screens taken to a fixed voltage
of about
+270V
will

give much better linearity than triode or UL screen taps. Such
an
arrangement has

voltage NFB applied relationships between grid and cathode and
between
screen and

cathode. 20% CFB with
fixed screen supply voltage Eg2, is the equivalent of having

20% UL taps but with about
10dB of series voltage NFB between cathode and grid

when a 4.7k load is used. Thus THD < 2% THD
at 9watts. This oversimplifies what

is occurring in the tube, because there are two
paths of NFB, one via the screen to

cathode interface and the other via the grid cathode
interface. There is still some

odd numbered harmonic products in the THD but they are much
reduced
below the

levels occurring with pure beam tetrode. The results using
EH6550 ( or
EH6CA7 )

in my SE 35 Watt amps with CFB indicated that more could be
achieved
with CFB

than
just UL screen taps and still have gain about equal to triodes.
The SE
35 Watt

amps
have remarkably low THD and low Rout and with only very minimal
global

NFB. See my page
on SE35
amps.