SE OUTPUT STAGE
This page explores
single ended output stage properties with beam tetrodes and
The aim is to produce the best audio fidelity which includes :-
1. Maintaining high enough gain of output tubes with local NFB
while enabling drive
voltages at low at low distortion,
2. Minimizing any reliance on large amounts of global NFB,
3. Locally applied NFB within the output stage to reduce
distortion and Rout to very
low levels to give better performance than triode strapped
4. Maintaining unconditional
stability which means no possible combination of L, C
or R loads, or with no load at all can cause oscillations at any
5. Ensuring that stability is possible with critical damping
R&C networks to reduce gain
and open loop phase shift below 20Hz and above 20kHz.
6. Maintaining bandwidth of 10Hz to 65kHz with pure resistance
load at the -6dB output
voltage level, ie, 1/4 full rated power,
7. Ensuring that output power at -1dB below clipping is possible
without any stage
of the amp being overloaded of forced into grid current draw
between 20Hz and 35kHz.
Single ended output stages have been used for audio amps ever
since the first amplifiers
were used for radio and
audio frequencies after the invention of the triode in 1903.
Up until about 1960 nearly
everyone in the world listened to radios and TV sets which
had one tube devoted to powering a loudspeaker.
Push Pull operation could give
more power than most ppl ever needed, and were
initially more expensive to produce than using just one output
triode. But once beam
tetrode and pentodes were invented, PP amps became popular
because they could
operate in class AB with low bias current. Such PP amps required
less power from
batteries or mains so construction costs declined which suited
However, if a single
large tube or a number of paralleled tubes are used in pure
class A1 audio power, the sound produced may be better than many PP amps.
I'd rather have one KT88 in triode making 8 Watts than having
two 6V6 in PP
making 10 Watts in class AB.
Content of this page is based
Fig 1. Three most used basic SE amp stages, 13W
to 10W, SE pure tetrode,
SE Ultralinear, SE triode, 1 x KT88/6550.
Fig 2. 20W+ amp with SEUL with 2 x EL34, KT66,
KT88, KT90, KT120, etc.
Fig 3. The Equivalent Model of KT88 with g1 and
g2 inputs treated as current
This allows understanding of
operating properties of a KT88, and its Ra, gm g1,
gm g2, and to analyze all voltages and currents in all electrodes to
voltage gain, and effect of local NFB. The theory may be applied to all
including pure triodes which do not have a screen, such as 300B
Fig 4. 20W+ amp with CFB, with 2 x EL34, KT66,
KT88, KT90, KT120, etc, using
same SEUL OPT in Fig 2.
Fig 5. 36W SE amp with CFB with 3 x KT88 etc,
using SEUL OPT, 25% UL tap,
1k3 : 5r6 Z ratio.
This was designed to include
"choke sink" for cathode current, and choke in anode
feed for driver tube.
Probably nobody has
ever built an amp like this because they need to source
good quality chokes.
The THD and Rout is much lower than conventional SEUL amps.
Fig 6. 25W PSEUL + CFB
amp designed around the Hammond 1640SEA output
transformer with mosfet CCS at KT88 cathodes,
Fig 7. 25W PSEUL + CFB amp designed around
transformer with choke at KT88 cathodes,
8. SE CFB output stage and SEUL output stages with OPT
with 3 windings.
Fig 9. Choke Feed SEUL
and SET output stages.
Fig 10. Choke Feed SET
amp with 845.
Fig 11. Choke Feed SEUL
with floating B+ supply.
Fig 12. Choke Feed SET
with floating B+ supply.
There are a number or ways to arrange output tubes, OPTs, and
and NFB. It is impossible to always consider the output stage
and without the interaction of the driver and input stages and
the NFB loops.
BASIC SINGLE ENDED AMPS.
Everyone should be familiar
with the basic class A working
of a single vacuum tube
( or a number of them paralleled ) in a circuit with tubes, OPT, and
PSU all in series,
with an idle current which is varied between zero idle current and twice idle
maximum audio output power.
The operation of the tube will
always remain predictable in terms of applied rail
voltages and applied signals to all electrodes. The Ea, Eg2, Eg1 and Ia may all
be varied within a range to give ideal working with a chosen
Fig 1. Basic configurations of SE amp stages.
The main 3 varieties of "Single
Ended" output stages used in many simple tube
amps are shown here.
I am using a modern Svetlana KT88 to show what is possible but
of KT88 or 6550 could be used similarly.
The general principles apply
to the range of power tubes now readily available,
could be KT120,
KT88, 6550, KT66, 6L6GC, 5881, 807, 6V6, 6CM5, 6CA7,
EL34, EL33, EL84, EL86, GU50, 13E1,etc, etc.
SE Beam Tetrode or
See Far Left side of Fig 1.
The use of single ended amps with one beam tetrode or power
pentode offers high class
A efficiency of up to 45% but with high THD with a dirty mix of
both even and odd
numbered harmonic products from low levels to clipping level.
The THD mix of H varies
much with load value. The IMD is appallingly high and Rout is
high and to make such a
tube configured here work far better, at least 20dB of GNFB must
be used from OPT
sec back to an input port of an input tube, usually its cathode.
Many such amps have
been built with an output stage like the one shown and all seem
to me to be rather
damned awful. Put it this way, if you use an EL84 to make 4
Watts in pure SE pentode
mode, and with 20dB GNFB, then its sound just is not as good as
using a 2A3 to make
4Watts with say 12dB GNFB. Millions of AM radios made before
1955 had a lone 6V6
for the audio amp and no GNFB. Although the sound of music was
awful the set was
cheap because most ppl were poor, and you could understand the
nightly news and
cricket scores, and be manipulated by politicians and
Beam or pentode tubes are easily driven because gain is highest,
and drive for class
A1 never involves class A2, where grid voltage goes positive
during input cycles so
grid current is then drawn, with huge reduction in grid input
SEUL Beam Tetrode or Pentode.
See Center of
The use of a tap along the anode primary
winding was rarely ever used by any
manufacturer because it cost more money to put the tap there,
and CEOs hated
innovation because it always cost money, and CEOs don't always
agree with the
marketability of innovation. Besides, the idea of using an
"Ultralinear" tap was not
invented until maybe just after WW2. With a tap at 40% to 65% of
anode primary turns
the beam tetrodes or pentodes acquire triode like harmonic
products, less IMD, while
maintaining anode efficiency at over 40%. Gain is less than pure
beam / pentode and
grid drive remains free of grid current. There is nothing
"ultra" about the linearity of
having a screen tap for "partial triode" operation, but in
general the UL connection
makes the tube work very much better than otherwise, and
sound heard will verify it. Some GNFB is still needed, usually
SET - Single Ended Triode.
See Far Right of Fig 1.
In the early days of electronics only one triode was used for
The idea remains well respected as we all see with 45, 2A3,
300B, 211, 845,
GM70 etc, for SET triode power from 2Watts to 22Watts.
With real triodes listed, there was no screen grid connection
shown above, and old
OPTs never had a screen tap on OPT. All the common varieties of
beam and pentode
power tubes may be configured as a triode by connecting screen
The screen then carries ALL the anode signal. This exerts
electrostatic control of the
electron stream in similar manner to the control grid. The gm of
the screen g2 is usually
between 1/20 to 1/6 of the gm of g1 control grid. The screen
performs the task which
the anode in a real triode performs. This function is an
application of local NFB.
However, the function is not a linear function, so despite the
high amount of inherent
NFB in triodes or triode strapped multi grid tubes, there is
usually about 5% THD at
full Po on the one load value which gives maximum Po. The
efficiency of triodes or
triode connected tubes is between 15% to 33%, depending on the
conditions. Higher efficiency of up to 40% may be gained by
using a direct coupled
cathode follower to drive the triode grid for class A2
operation. Usually class A2 is
more bother than its worth and if more triode Po is wanted, use
more triodes and
stay in class A1. A renowned exception is Audio Note "Ongaku"
amp with 211,
which has a 6SN7 cathode to drive its grid. Not all beam and
pentode OP tubes
can be easily driven in Class A2 such as EL34. Grid current is
just too high.
It is difficult to make a bad
sounding triode amplifier providing no tube
is driven into any overloaded operation. Triode mode is the
"safest bet to good music".
The Ra of triodes is usually much less than the anode load
driven. At low levels
THD and IMD can remain quite low enough so GNFB may not be
THD at all levels is mainly 2H. Gain of triodes is perhaps 1/3
that of beam or pentode
tubes and the drive voltage level may be quite high, up to about
110Vrms with 845.
But providing enough drive is never difficult by use of another
SE triode driver tube.
The 2H of the driver cancels part of the 2H produced in the
So at all levels up to clipping, some SET amps produce
surprisingly low 2H which
gives less objectionable THD and IMD than from a PP amp of the
Few radio sets
ever had slightly larger output tubes than 6V6 or 6F6 such as
807, 6L6, or later EL34, KT88, 6550. But fabulous sound quality
could be had from
these tubes strapped as triodes. In fact, in many radios I have
re-wired, it is always
possible to get at least 3Watts from a single EL34 strapped in triode and power is
the same as one 6V6, and fidelity with trioded EL34 is far superior to
The above output stages are all
based around Russian Svetlana KT88 and similar
Russian EH 6550, KT88. Tubes such as KT90EH and KT120EH may
always be used
with slightly higher Ia, cathode bias Rk to suit, and a lower RLa load to be used
obtaining the same Va, hence SE Po up to 15Watts is possible. But an OPT meant for
4k2 : 4r,8r,16r and Ia = 100mA max cannot be pushed by
increasing Iadc much beyond
Notice that for the same B+
supply, and similar Ea and Ia, the triode load value is lower
than for tetrode or UL. See my pages on loadmatching to SE triodes and pentodes
work out operating conditions different to the above.
To get better triode
performance the B+ usually needs to be say 15% higher than for
the same tube with UL. Then the same OPT can best be used
In the above example for SE Triode, the OPT can be 4k2 :
If the B+ is raised to +490Vdc,
Ia +Ig2 reduced to 70mAdc, EK = +52V, and Pda+g2
remains 30W. Triode Po then becomes 10W. KT88 or 6550 then give very
performance to 300B.
Fig 2 shows a very ordinary
20Watt Ultralinear output stage using a pair of EL34
tubes in parallel, and based on using an available Hammond
1627SEA OPT with
ZR = 2k5 : 4r, 8r, 16r.
My comments below are based on
using EL34 or 6CA7, but parallel pairs of other
tubes such as KT88, 6550 can be used.
There is no NFB applied in the
EL34 output stage except for the local "ultralinear
screen tap" which is commonly not considered local NFB, ( although it actually IS NFB
The UL tap allows the high power of class A1 pure pentode but
with triode like
harmonic products, ie,
less odd number H and more even number H. Ra is
reduced greatly so a better damping factor is possible than for
pure pentode / tetrode.
All triodes or triode strapped
pentodes or beam tetrodes have internal NFB by
means of the field effect of the anode interacting with the field effect of
grid to give a resultant effect on the electron stream from
cathode to anode.
See my equivalent models of
multigrid tube basic operation with Ra in shunt
with g1 and g2 current generators.
For multi-grid tubes such as
EL34, KT88, having Eg2 a fixed Vdc hugely reduces
the signal field effect of anode upon Ia current. The screen g2
may be connected
to a source of signal voltage which is a fraction of the anode voltage and this
has an opposing action to whatever the g1 signal voltage tries
A diagram about basic tetrode
of pentode operation may explain more lest I
completely confuse everyone.
Please remember that the beam
tetrode has no suppressor grid as found in the
pentode, but has beam forming plates connected to the cathode internally
function like a suppressor grid g3. There is no need for mention
suppressor grid or beam
formers again because the relevant operation facts
are concerned with control grid g1 and screen grid g2.
If you can follow and apply what you find in my Fig 3 diagram,
you will find it
useful to analyze possible signal voltages and currents like our ancestors did,
and all without PCs and simulation programs.
Understanding models frees you from any need for a PC. But
ancestors who used a slide rule you may use a $5 pocket
Fig 3 above shows an equivalent
model for what is happening with 1 x KT88.
Other tubes such as 6550, KT90,
KT120, 6L6GC, 5881, KT66, 6V6, EL34, EL84, etc,
will show similar basic operation, and by understanding the model you
any output stage or design one. But you MUST start by knowing
the g1 gm and g2 gm,
and the Ra for the tube operating conditions of idle values
of Ea Vdc and Ia dc..
In the above Fig 3, the figures
for signal current and voltage amplitudes have been
prepared from measurements of transconductance of g1 and g2
samples of tubes under the conditions listed for Ea, Ia, Eg2.
The Fig 3 figures above don't
apply to all output tubes but will agree with curves
shown for Svetlana KT88, and other Russian makes of KT88 and 6550.
Testing tubes in a real world
amplifier isn't difficult after you learn what you are
doing which may take days of practice.
Hum voltages present in output
stages may spoil the attempts to measure small
signal voltages in an amp. To avoid noise interfering with
there is adequate filtering of the B+, and if not, apply much
larger filter caps and
perhaps a choke between rectifier and the existing B+ filter
caps. This is very
necessary in SE amps where there is no common mode rejection of noise in B+ rails.
But let us suppose you have an
existing SE amplifier with a multigrid output tube,
and you have well filtered B+ rails for anode, and g2 and input
To determine g1 gm :-
g1 gm is the g1 transconductance which is the ability of the g1
grid to vary
the electron flow and is expressed in mA / Volt.
1. Connect 100r resistance between anode and OPT anode
2. Connect wire shunt link with alligator clips
from OPT anode connection and
B+ connection, to short circuit the primary of OPT.
3. Make sure the screen g2 has the intended
fixed supply voltage.
4. Disconnect any global NFB and output speaker
5. Connect audio signal gene to amp input and
use a 1kHz sine wave and
increase until there is 1Vrms at the output tube g1 grid.
6. Measure the signal voltage across the 100r
and have a CRO connected
to anode to make sure distortion is less than 2%.
Suppose you measure 0.64Vrms (
Vac on DMM ) across 100r.
7. Calculate g1 Gm. The current in 100r = 0.64V
/ 100r = 6.4mAac.
This means the 1V change at g1
produces 6.4 mA change at anode so
gm = 6.4mA / Volt. Simple?
The anode signal voltage
variation will cause virtually no effect on the gm
measurement because the gain between g1 and anode is negligible
with a very
low RLa of only 100r. Hence there is no action of internal NFB
within the tube.
8. For triode connected tubes, the same set up
works OK but g2 is connected
to anode, and the gm measured will vary slightly, because g2 signal
generated in addition to the anode current. Triode Ra may also
The load of 100r is a
near vertical load line is drawn upon data sheet Ra
curves for KT88. We wish to understand the basic current change behaviour
caused by grid voltage change.
Measuring g2 gm :-
The g2 gm is the screen g2 transconductance which is the
ability of the g2 grid to vary
the electron flow and is expressed in mA / Volt.
1. The above set up is used with 100r anode to
B+ load, and OPT primary
shunted by a wire link.
2. Ground g1 with 2uF so there is no g1 signal
possible, but the wanted g1
Vdc bias is undisturbed.
3. Disconnect g2 screen is from wherever it is
normally connected, and insert
1k0 in series with g2
and original g2 B+ supply, so that g2 retains its Vdc bias supply.
4. Connect a signal gene between 0V and g2 but
use a 2uF DC blocking
cap to enable up to 5Vrms at 1kHz to be applied to the g2.
5. Measure the signal voltage across the 100r
anode to B+ resistor.
Suppose you measure 0.45Vrms at
100r. It means 5V at g2 produces 0.0045
amps of anode current.
6. Calculate g2 gm = 0.0045A / 5.0V = 0.0009A/V
Measuring Ra :-
Ra is the internal dynamic
resistance of the tube when operating in pure beam
tetrode or pentode, and it always exists in the current gene
model and it varies
depending on Ia and to Ea.
Ra is a difficult parameter to measure accurately for beam and pentode tubes
because it is a high number of ohms. Ra will be easier to
if there is a UL tap
or triode connection, and then the Ra is the beam or pentode
Ra IN PARALLEL with the effect of the g2 current generator. When
the g2 signal
voltage remains the same as the cathode voltage ( with a cap
bypass between g2
and k ), then the g2 generator has infinite impedance between
the anode and k.
1. To attempt to measure pure tetrode or pentode
Ra, disconnect all shunts across
OPT winding and have g2 taken to the wanted B+ fixed voltage.
2. Remove any R&C zobel networks across the
OPT secondary or primary.
Remove any secondary load or global NFB connected.
3. Make sure cathode is well bypassed to 0V.
4. Connect 1k0 between anode and a signal gene
with DC blocking cap = 1uF.
((( This can be tricky if you have a solid state signal gene because such test
gear can be fried to a useless crisp if their outputs are
subject to external voltages
exceeding say +/-20V. So
you need to have adequate protection measures on
inputs and outputs of ALL TEST GEAR !!!!!
Using a tubed cathode follower in a separate box is good practice for connections
between delicate SS test gear and tube gear.)))
5. Ground g1 of power tube with 2uF cap and
ensure g2 is grounded without a
series screen stopper resistance.
6. Apply 10Vrms at 1kHz to input side of 1k0
resistance from signal gene.
Measure the anode to 0V Va-c, and record this.
7. Measure the Vac across the 1k0.
Suppose you measure 0.4Vac
across 1k0. Then current flow from anode to
0V = 0.4V / 1k0 = 0.4mAac.
Suppose you measure Va-c to 0V
8. Calculate Ra, dynamic
anode resistance between anode and cathode
= 9.6Vac / 0.0004Amps ac = 24,000 ohms.
You will quickly realize that
using the published data sheet values for gm and Ra
will lead you to gross mistakes in calculations of what is going on in your tube.
A KT88 with Ia = 60mAdc may have g1 gm = 6.4mA/V, but the data
us g1 gm = 11mA/V at Ia = 140mAdc. Nobody would ever have a KT88
Let us suppose the OPT
Lp = 25H, and at 1 kHz this has inductive reactance XLp
= 157,000 ohms, with perhaps 400pF of shunt C which makes XC = 397,000 ohms.
The XLp, and XCsh will be found to have a negligible loading
effect at 1kHz
because usually you
will find Ra is always below 50k for most large multigrid output
tubes such as KT88, KT66, EL34. So you may ignore the current flow
gene through primary inductance and any shunt capacitance. ( But
of course there
are some very badly made
amps around and you can't
assume XLP and XC will be negligible. So measure these
parameters as well.
Always avoid making any stupid
Measuring triode Ra in the
manner above but with g2 tied to anode, expect Ra to
be 1k1 for KT88.
To measure the amplification factor µ.
amplification factor µ is the calculation of gm x Ra. It
is in fact the voltage gain
of any tube where a grid causes anode voltage change without any
This means the anode load = infinite ohms, and the loadline for
an infinite load
is a horizontal line across the data Ra curve sheets for Ea and
The µ of beam and pentode tubes is the most constant
parameter, with large
variations of gm and Ra. Triode operation makes µ much
more constant for
wide Va swings.
1. Set up the amp as it
would be for normal operation, but with no global NFB loop,
no loads or zobel networks.
2. Apply 1kHz sine wave to output tube grid g1
via 2uF cap. Use DMM or hi-Z
input volt meter to measure Va to 0V.
3. Connect CRO to monitor the Va signal
4. Increase g1 signal to obtain Va 50Vrms. All
output tubes should be able to
achieve this much voltage with no load, and with THD < 5% for multigrids
less than 1% for triodes.
5. Measure the grid input voltage.
Suppose you measure 0.312Vac.
6. Calculate voltage
gain and µ without any load = Va / Vg = 50 / 0.312 = 160.
It is just a number, no units. It may be considered a negative
a negative going voltage at g1 or g2 produces a positive going
voltage at anode.
The voltage gain without any signal load current = amplification
7. Verify measurements and calculations. For all
tubes and for any g1 or g2 input,
transconductance, gm, = µ / Ra.
If we knew gm, and Ra, we could
calculate µ. From the above examples,
g1 gm = 6.4mA/V and Ra = 24k, so µ for g1 = 0.0064 x 24,000 = 153.6, close
enough to the 160.
The Ra in Fig 3 is always present as a shunt resistance
in Fig 3 and is in
parallel with the effect of the g2 current generator if there
is a signal voltage
applied between g2 and cathode. Thus the g2 generator can become equivalent
to a resistance.
Consider Fig 3 with an OPT set up with a 25% UL
tap for g2 so while there is
+100Va-c+ at anode, +25Vg2-c appears at g2.
The +V at g2 increases current
flow. it is calculated = Vg2 x g2 gm
= 25 x 0.9mA/V = 22.5mA. If you went looking for the 22.5mA, you would not find
because it is a "useful imagined current" that exists in a mathematical model to
help explain what we do measure in a real circuit.
Consider Fig 3 without the 4k0
RLa anode load. There would still be the imaginary
22.5mA with Va = 100Vac.
But there would be no load
current. The Ra of 25k is there, so the current in Ra
= 100V / 25k = 4mA.
Total Ia model current change =
22.5mA + 4mA = 26.5mA.
The effective Ra = Va / Ia = 100V / 0.0265A = 3,773 ohms.
The portion of this resistance
due solely to g2 generator = 100V / 0.0225A = 4,444 ohms.
The effective Ra' for where a
screen signal exists = Ra // ( 1 / UL fraction x g2 gm ),
where Ra is that for pure tetrode or pentode.
Example :- Effective Ra', 25% UL for KT88 = 25k
// ( 1 / 0.25 x 0.0009 )
= 25,000 // 4,444 = 3,773 ohms.
Example :- Effective Ra', 50%
UL for KT88 = 25k // ( 1 / 0.5 x 0.0009 ) = 25,000 // 2,222
= 2,040 ohms.
Effective Ra' for trioded KT88
= 25k // ( 1 / 1.0 x 0.0009 ) = 25k // 1,111 = 1,064 ohms.
From the calculation of Ra' for
any % of UL, we can also calculate the UL g1 µ.
It will always be be less than pure beam tetrode / pentode
µ and more that triode µ.
µg1 = g1 gm x Ra, and if
25% UL Ra = 3,773r then g1 µ = 0.0064A/V x 3,773
The screen has its own transconductance value g2 gm, based on
having no g1 signal.
The pure beam or pentode Ra of the tube remains the same if the
screen is used
alone for Ia change and Va change.
µg2 = gm g2 x Ra = 0.0009A/V x 24,000 ohms = 21.6.
Screen gm is so much lower than g1 gm that screen drive is
seldom ever used,
so pure screen driven output tubes are not discussed here. Feel
free to experiment
to know more about screen drive properties. But try as you may,
you will find the
model in Fig 3 very useful if you understand the concept of the
basic input voltage
controlled current source which has infinite resistance looking
into its output at the
top of the two circles on the drawing.
The voltage gain, A, with a load for UL can also
be calculated A = µ x RL / ( RL + Ra ).
For 25% UL, in this case, A =
24.15 x 4,000 / 7,773 = 12.42
For 100V into 4k0 load, the
input voltage = 100 / 12.42 = 8.05Vg-c.
The only confusing thing
remaining is the + / - sine for Vac. All tubes in "common
mode like in Fig 3 are
inverting, ie, a + input voltage causes a - output voltage.
So as said before, in effect, µ is negative, -, so
-8.05Vg-c produces +100Va-c.
The damping factor is calculated DF = RLa / Ra.
Typical values for a single KT88 without
any CFB or global NFB and as shown in Fig 1,2,3, and for RLa =
Pure tetrode, DF = 4,000 /
25,000 = 0.16 = very poor. 20dB NFB needed.
25% UL, DF = 4,000 / 3,773 =
1.06, much better, but 15dB NFB needed.
50% UL, DF = 4,000, 2,040 =
1.96, better, but 12dB NFB needed.
Triode, ( 100% UL ), DF = 4,000
/ 1,064 = 3.76, 12dB NFB needed, but some can't hear
any difference with NFB.
CFB over 10%, fixed Eg2, DF generally as high as triode or
The maximum THD of SE output stages without
local FB is highest for pure
pentode or tetrode and can be 15% for where the RLa is that used to
maximum possible Po and at 1dB below clipping. Many even and odd
are present at all levels.
For class A1, the amount of UL % can be up to 70% for most
and the Po max without grid current is nearly the same as pure
tetrode or pentode.
Odd H are much reduced and
THD will be around 7% of mainly 2H. But where
the UL connected tube produces the same Po as lesser maximum for
operation, THD may be less than the triode.
The less THD one has before NFB is applied, the better is the
The FB network allows a fraction of the output distortion H to
be applied to an
amp's FB input. The input tubes and output tubes are not
and the signals fed back create distortion H, so
intermodulation, or "IMD" H
are generated which are the sum and difference between any two
If you have 3H fed back with fundamental test tone of 1H, then
the IMD products
will be 3+1 = 4H and 3-1 = 2H. Their level will depend on the
non-linearity of the
So by means of intermodulation, H products appear at the output
not present when the amp was tested without NFB. It is a very
where one has a pure beam tetrode making 10% THD and where only
global NFB is applied, and one will find some reduction of 2H
but increase of
3H and 5H and perhaps other intermodulation IMD H that were not
by the beam tetrode. Where the signal is music with perhaps 20
present, they all react to modulate each other even without any
and when NFB is applied in small amounts the fidelity betterment
is less than
expected if the THD for a signal tone was 10% to begin with.
With 10% before
FB and with only 8dB NFB applied, the reduction of THD and IMD
will not be 8dB,
and the sound may not seem any better. But a tube making 10% THD
NFB will measure 1% THD with 20dB NFB.
The "second order" IMD H products may be less than 1%.
But as the H number rises, their audibility increases by factor
of N squared / 4,
where N is the H number. Hence it takes very little 7H to make
an amp bad where
the input signal = 100Hz, and the 7H = 700Hz. If you have 100Hz
+ 500Hz present
at high levels then you get IMD H at 400Hz and 600Hz and if
these H are NOT
musical tones in the music scale their presence lessens
perception of fidelity
because they are not in harmony with music.
The THD if considered alone make little difference to perceived
fidelity if it is
below a few % because most musical notes have many H already
altering their levels by a few % makes little difference. But
where the THD
does measure above 0.5% the perceived fidelity begins to suffer
because of the
INEVITABLE production of IMD as a result of the non linearity
which we measure
by using a single pure sine wave input to assess THD and the
resulting H spectra.
With music, if you could listen to all the IMD products and THD
without the original undistorted music, you would hear a bad
harmonious noise rising and falling in level according to the
It is disgusting junk that poisons our enjoyment.
Applied FB should effective over a wide F range, and open loop
gain of the amp
without any FB should be wide as possible, and phase shift
10Hz and 30kHz. What is needed to minimize production of IMD
tubes is to get them to be linear by NFB applied locally without
NFB around 3 amp stages. The use of CFB and UL taps convert non
tetrodes and pentodes to being quite different in that the
initial spectral complexity
is reduced and simplified more than NFB theory would predict.
The amount of global NFB applied can then be quite low without
"extra H" IMD products providing we make the input and driver
In most tube amps, THD produced by driver tubes is much less
than output tubes.
So multigrid tubes benefit with UL taps, and more with triode
and even more with CFB with a tertiary winding on OPT. Adjusting
level applied between screen and cathode is very important along
setting the screen grid Vdc, Eg2, to be just right.
Usually, although most beam tetrodes and pentodes produce
THD levels of up to 15% in pure beam / pentode mode at just
they can be cajoled into producing less than 2% with local NFB.
It means their
gain is reduced from say 20 to 3, which means that instead of
drive we may need 70Vrms, but this is easy, and at low THD below
Is it possible to use the same tube and OPT as in Fig 1
local CFB in the OP stage?
SE amps with CFB will normally
have the OPT primary winding divided into
2 windings, with between 70% and 90% of the turns in "the anode circuit" and
the remaining 30% to 10% of turns in "the cathode circuit".
The signal Ia flow and Idc flow is equal in both windings except for
minor difference occurs with dc and ac screen current.
An example is at
There is no strict ideal ratio
of anode turns to cathode turns and I have
used up to 2:1
successfully in my SE32 amps, 2012 version, with 13E1 tube,
ie, anode turns
= 66.7% of total, cathode turns = 33.3%.
Hardly anyone has the patience,
time, money or skill to make their own well
designed OPT to suit multiple paralleled output tubes with CFB windings.
Few companies produce any OPTs with any CFB windings as a stock
buy off the shelf. So OPTs for CFB use usually have to be custom
a far higher price than "normal" stock items, and you wait
months and deal
with bullshit artists. There is little doubt my efforts to make
the SE35 and the
SE32 creates sound quality to keep amp owners happy ( 2014 ).
But now let us question Fig 2
Could the standard off the shelf Hammond OPT be used to provide
cathode feedback in the output stage, despite the fact there
is only ONE
"anode" primary winding?
Is it possible to "build the tubes around the OPT"?
Here is my schematic......
The above Fig 4 uses a floating
B+ anode supply which in effect works like a 400V
battery between the tube anodes and the anode connection of the OPT
The UL tap is connected to 0V.
The normal connection for B+ is
taken to cathodes and their biasing R&C networks.
This arrangement has
anode Idc and signal Iac flowing around the whole series circuit
formed by tubes, primary winding, and PSU, just like any
But here I have the whole primary at 0Vdc potential with UL tap
taken to 0V and the
non grounded power supply is moved to between tube anodes and
of OPT. The tubes are operating in exactly the same manner as
they would in Fig 2
and relative signal voltages are the same.
There is a total of +229Vac
across all turns of the primary. But -91Vac between
cathodes and 0V, and +138Vac between anodes and 0V. The -91Vac and +138Vac
have opposite phases.
To make this this happen,
-22Vac must be applied between grid and cathode,
and this must be added to to the cathode voltage of -91V t give
((( Some would ask, if you add -22V to -91V, don't you get -69V?
All I can say to them is that they need to study more, observe
more, measure more,
build more, and then finally what I am saying might just make
A normal UL stage as in Fig 2 has the screens taken to a UL tap
at same +Vdc as
the anode. In Fig 4 above the relative UL signal voltages remain
so screens must be be connected to the UL tap. But the screens
must be at a high
+Vdc potential so there must be an additional fixed B+ supply
rail applied between
screens and UL tap. This B+ rail supply is just a normal
grounded B+ supply
which will be also used for the input and driver stages.
The final result has an output
stage with quite low overall gain compared to the
normal UL OPT stage. The conventional UL stage has
voltage gain of about
10, but the CFB stage has gain of only 2, with 113Vac needed to produce 229Vac
across the primary of OPT. There is a very worthwhile advantage because the
effective output resistance of each EL34 reduces to about 290
The pair of tubes makes the total tube Rout = 145r, and damping
RLa = 2k5 becomes 2,500 / 145 = 17. The reduction of gain and distortion
OPT stage is a factor = 22 / 113 = 0.195. This means that the typical UL THD
of say 6% at just below clipping is reduced to about 1.2%.
But there's a price to be paid
for the output tube performance betterment,
and the drive voltage applied to the output stage must be 5 times
greater, and this
must be created without much distortion. My usual method is to use an EL34 in
triode mode with choke+R for the dc feed to easily make up to
140Vac at less
than 1.5% THD. An
example of this is at my SE32 amp at
There is an additional benefit
not easily recognized. The rate of 2H production
in the output stage is similar to the rate of 2H produced by the
driving EL34 triode.
Therefore there may be considerable cancellation of the 2H produced in the output
stage. This cancellation is load dependent, most reduction of
THD by 2H
cancellation occurs in
the middle range of load values, and the maximum
cancellation causes THD to become as low as that found in a well designed
class A1 push pull amp.
There is no absolute need to
have a floating PSU B+ supply for the OP tubes
as in Fig 4. A normal grounded B+ supply may be used as I have it
Fig 5 shows a
complete amp using a conventional SEUL OPT with just one tapped
primary winding. I
designed this schematic for my friend Vali in Romania.
He wanted to make a 2 channel chassis and using 3 x KT88 for
He found a supplier for SE OPT
for 30W+ into 1k8 : 8r0, with UL taps at 25%
The properties of the OPT were analyzed and after some calculations, I thought
3 x KT88 could make 36Watts into primary load of 1,300, with sec
Thus he ought to get at
least 30W at the 5r8 output load. The amp would
tolerate all loads between 3r0 and 20r0.
But I have since made a few improvements to the schematic and I
anyone else wanting to make such an amp may not be able to
source the same
OPT which my friend Vali found. So therefore I have drawn the
UL taps at 25% and 50%, and with tapped secondaries for speakers
4r, 8r, or 16r.
In Fig 5, there is NO FLOATING
B+ anode supply as shown for Fig 4, and there
is just ONE B+ supply used for all B+ rails for the amp.
The B+ apply of +410Vdc is
applied to the 25% UL tap so anode current flows
from B+ to the normal anode connection. The KT88 anodes have 75%
total primary voltage, in this case 162Vrms.
Having Idc flow in 75%
of the primary winding means the DC core magnetization
Bdc is only 75% of what it would be if the DC flowed in all turns. Instead of say
Bdc = 0.7 Tesla, it is 0.53 Tesla.
So Bac can be 133%
higher at 0.93 Tesla, instead of 0.7 Tesla. So the frequency
of core saturation will be 0.75 x original design Fsat, so if Fsat was say
25Hz for full Po level it would be reduced to 19Hz, a
OR, one could have Fsat at 25Hz but at a considerably higher
In other words, with the same Idc in fewer primary turns the
output power can be
The signal current produced by KT88 flows in ALL the primary
those without any Idc. The original B+ connection on the primary
is taken to the
KT88 cathodes via C20, 470uF. So where does the Idc flow from
It flows down to 0V and back to PSU via L2, 5H choke.
In this case, the OPT had a 25%
UL tap which is an ideal % for deriving CFB.
Some available OPTs may have UL taps at somewhere between 25%
The 50% UL tap is in fact the CT of the primary winding.
If the 50% UL tap was used for B+, then in this case the anode
signal would become
108Vrms- and cathode signal would be 108Vrms+, and the the
drive signal would be about 20Vrms+ so the EL34 in triode mode
would need to
make 128Vrms+ which is getting a bit high to achieve at low THD
But in my SE32 with 13E1 I have an EL34 in triode which must
100Vrms to drive a tube with 33% CFB and it sounds just fine
The KT88 in fig 5 are working with relative electrode voltages
identical to a
conventional UL amp with normal 25% UL taps. The Fig 5 circuit
has 25% of
the Va-k fed back to the cathode. This cathode signal is in series with the input
grid signal. The amplitude of the cathode feedback is sufficient
to be a very
effective amount of local NFB which reduces the THD by a factor of more than
1/4, and reduces output resistance of the stage to about 1/2
that of triodes. But
maximum drive voltage needed by the output stage is still at a
moderate level of
72Vrms and very easily produced
by an EL84 strapped in triode mode.
The L2 5H and C20 470uF
form an LC filter with a pole at 3.3Hz, but I believe this
F is so low and that cathode output resistance is so low that
resonant effects of the
C&L are suppressed, ie, damped, and LF stability should not
L2 choke should be at least
5H, and with Rw about 50r max. However, if Rw was
100r, then the top of the choke would be at +26.6Vdc and the
cathodes would be
at +60.3Vdc, so the B+ supply would need to be raised from
+410Vdc to +424Vdc.
Amps like this should always be built with a few taps 15Vrms
apart at the end of the
winding to allow B+ to be varied depending on winding
resistances and to
accommodate changes to the total load value seen by tubes to
best suit an OPT.
L2 choke shunts the cathodes to
0V with its increasing low reactance as frequency
becomes lower. This may seem to reduce the NFB and increase KT88. The
is also increased by C20 whose reactance increases as F gets
lower. But at very
low F of say 3Hz, the OPT primary reactance shunts the tubes to
an extent that
whatever weird behaviour occurs with C20 and L2, it is of no
The C20 should be at least 470uF rated for 450Vwkg, or a pair
should be bypassed with 2uF to shunt the HF impedance of the
The choke will saturate
at some low F below 14Hz. It needs to be designed to
take up to 100Vrms with 265mAdc present without core saturation
many "off the shelf chokes" may not fulfill the condition, so
perhaps a pair of
smaller 2.5H chokes in series would be better.
At 20Hz, the L2 5H has XL
= 630r, and C20 XC = 17r. The inductance of the
whole OPT primary should be no less than 10H. The Lp acts to shunt cathodes
to anodes and reduce the Va-k. as F goes lower. The CFB tries to
the Va to k with NFB action, but the tubes saturate at 20Hz at
full Po so
output voltage is limited and the L2 and C20 have little effect
on overload behaviour.
This SE amp, like all others, should have an C&R HPF at
input plus LF gain shelving
R&C network after V1 to reduce the open loop gain to very
low levels at below 5Hz.
The use of a choke in Fig
5 is in effect a "cathode current choke sink" of DC from
KT88 cathodes to 0V.
It is a similar technique to
using the well known ( but seldom used ) choke feed from
B+ to the anodes with cap coupling of anodes to OPT. It is also know as
The choke is feeding Idc to anodes and the choke inductance is
in parallel with
capacitor coupled primary inductance. In Fig 5, the L2 5H is in
parallel with the
25% of turns of the OPT, and inductance of 25% of turns will be
so the extra L2 inductance has negligible extra loading effects
at very low F.
The choke L2 is able to do some of the function of an air gapped OPT and therefore
get more audio power from the OPT.
It also would be possible to
have a CCS ( constant current sink ) using solid state
and all similar to what I have in Fig 6 below.
But the SS devices must be arranged very carefully because the +
/ - cathode V swing
means the cathodes go to a negative peak voltage well below 0V
so there must be a
suitably designed negative voltage rail.
If the 36W CFB amp in
Fig 5 seems like too much trouble, or you cannot source
the esoteric rarely ever available well wound OPT, then maybe
will sound well.....
Fig 6 shows a
"nearly conventional" SEUL amp and a Hammond 1640SEA with
40% UL tap.
The specifications for the Hammond 1640 tranny has P : S ratio
1,250r : 4r, 8r, 16r.
The secondary is one winding for 16r and 8r is a tap at 70% of
turns and the 4r is
at 50%, ie, the secondary center tap.
The specified maximum Idc is 200mAdc. This means that peak
current change in
class A = +/- 200mA peak.
So maximum Po can be calculated = 0.5 ( 0.2 x 0.2 x 1,250 ) =
This is a little bit less than what Hammond say, as it is a
30Watt rated tranny.
But we can forgive them this minor discrepancy.
With 25Watts, primary signal voltage = 176.7Vrms, and there is
Now 20Vrms is a large enough voltage to be usable for cathode
we wished because it is 11.36% of primary signal voltage.
Anything over 10% is
useful. If we had just one KT88 producing 8.3Watts with OPT for
3,750r : 16r,
the primary voltage would be the same 176Vrms, but 16r sec would
1.5Vrms, only 6.5% of primary voltage which is an ineffective
amount of CFB.
The normal UL tap and GNFB loop would suffice.
What would happen if we used the 1640 secondary as a CFB winding
The P:S turn ratio will change so that the transformer P:S ratio
= 1,549r : 16r,
because now the signal current of tubes flows in secondary
This current is much smaller than the speaker current so the
thicker sec turns can
With primary load at 1,549r, it suits the use of 3 x KT88 with
each tube seeing
4,647r, a very nice load for KT88 operating very comfortably
with Ea at +310V
and Ia at 60mA. EL34 would also work very well. Pda + Pg2 per
= 18.6W + 1.55W = 20.15W.
The Idc total for 3 KT88 is 195mAdc, nearly Hammond's maximum
If the 195mAdc flows in 16r winding which may have Rw = 0.8r,
the Vdc across
winding is about 0.16Vdc, and be applied to a speaker, unless a
DC blocking cap
is used, which we will not to because there is insufficient Vdc
to polarize a large
value electrolytic cap. In addition, the Idc in sec raises the
Bdc of the core which
will saturate at a higher frequency. So there's two good reasons
NOT to have
the tube Idc flowing in the sec. But the tube signal current
"helps" things to happen.
Individual cathode biasing R&C networks are good for
paralleled tubes always
operating in class A. But we could have constant current sinks
instead of resistors.
The Fig 6 shows IRF610 used for each output tube cathode. But
other CCS with
Darlington pair connected bjts could be used because the base input
very high and voltage across R27, R31, R36 remain constant.
The cathode "bypass" caps are still required for each cathode,
but instead of being
taken to 0V they are taken to the top of the 16r secondary
The three CCS each have effective collector resistance > 50k
so having 3 in parallel
makes a very high cathode load which we may consider has
negligible effect in any
considerations. The three CCS act like a choke in Fig 5.
Notice that the 3 KT88 cathodes will settle at about +35Vdc.
Just exactly what Ek
will be may vary but I expect =35V, but samples may vary. Now
the cathodes have
20Vrms at clipping, so the V-swing is +/- 28V peak. The mosfet
drain connections will
also have +/- 28V peak, and the minimum voltage across the
mosfets should not
become less than about 10V. The Vg-s should never go negative.
I have the mosfets set up with gates at -14Vdc, and IRF610 data
says gate bias will
be -4Vdc approx at low Id. So sources should be at -10V, which
is -45V below Ek,
enabling +/- 28Vpk cathode swing. I've chosen to have the
required negative rail for
mosfets at -24Vdc. Its not too hard for anyone to make an
unregulated -24Vdc rail
and for 200mA. I should not have to spoon feed you such a
detail, and I just won't.
The Fig 6 has a large total amount of applied NFB in 4 "loops".
Each can be
considered, and perhaps discarded if deemed unnecessary.
It is the use of the UL tap from primary winding to supply KT88
with what is normal Ultra Linear screen FB used now since about
The effect of this connection makes the effective Ra of each
KT88 = approximately
3k0, a huge reduction from the pure beam tetrode Ra of 24k. Odd
number H are
reduced and spectra is brought closer to triode operation.
Local Cathode Feedback from OPT is used to reduce the Ra to less
KT88 triode value. The CFB is an external loop involving
linear working transformer
windings so that all even and odd number H are reduced by the
One could say nobody needs to have any more NFB in such an amp
grid input to KT88 requires 38Vrms, so we MUST use a driver tube
because a preamp
or PC sound card cannot make a linear 38Vrms.
Screen FB to EL34 is used from the OPT secondary. Instead of connecting
the EL34 to anode for simple triode operation, it is bypassed
with C12 to the CFB
from OPT secondary. One might ask why, well, it can be done, so
let us consider
the results. The EL34 has to make 38Vms at anode and although
would work just fine, we could have its screen fed with a signal
voltage that is less
than its anode signal so the EL34 is working in Ultra Linear
mode. There is a
convenient signal available and of the correct phase which is
the OPT secondary
signal of 20Vrms that is applied to KT88 cathodes. So EL34 has
52% UL operation.
The load for EL34 is the R17 and L1 in series which becomes high
for most of the audio band. The EL34 will be found to be very
nearly as linear as
triode strapped with such a load. Gain g1 to anode will be found
to be twice that of
triode. Now within the OPT secondary signal there will be
generated by KT88 AND those produced by the EL34. All these
are amplified x about 10 by the gain between g2 and anode to
create an "error" signal
that when applied to the KT88 grids they are then amplified to
oppose their own
production. The screen NFB is effectively about 20dB NFB,
although not a most
perfect form of FB, because the EL34 cannot ever provide less
THD than when
triode strapped and with the high RLa load value I have used.
But a typical EL34
in triode mode with RLa > 20Ra can make 100Vrms at 1% THD,
and at 38Vrms
perhaps 0.3%, mostly 2H and with 52% UL operation it may be
0.6%. But the KT88
may produce 2% THD, much more than the THD of EL34. The effect
of this screen
FB will reduce THD from 2% to about 0.4% at least, a huge
OK, so now the amp makes 25Watts at 0.4%, and we need only 2.2
Vrms input to
the EL34 grid so anyone may try all this without adding yet
another input tube V1.
At normal listening levels of 1 Watt, THD can be expected to be
0.07% which is
an excellent measured result for most SE amps, and I suggest ppl
try it out.
What other SE amp amplifier has just TWO active devices with
You say there are 4 tubes total, but the three KT88 act as one
because they are
paralleled, and they could be replaced by 13E1, or 4 x EL34, and
all I'm doing
is exploiting the screen properties of an EL34 so it can be both
input and driver.
This is possible because the ratio of g1 gm : g2 gm is quite
low. Put another
way, the g2 gm is a useful high value which can actually do a
lot. Suppose we
used a 6BX6/EF80 instead. The g1 gm could be 5mA/V, quite
useful, but g2 gm is only
0.083mA/V, and the fed back THD content is not amplified many
times. The EL34
performs far better, even though it is a power tube. Other
suitable tubes for screen
FB applications are 6CA7, EL84, EL86, possibly EL36 / 6CM5.
Global NFB from OPT sec to V1 cathode. V1 is a 12AU7 low µ
twin triode with µ = 17.
Gain with CCS anode supply via MJE350 is about 15x. The sound of
this tube is usually
just worth dying for. Here is has little to do, but the amount
of NFB = 14dB, so the 0.4%
I spoke of above is reduced by about 1/5 to 0.08%, say 0.1% at
25 Watts, and at 1Watt
THD may be 0.014%. Such figures are typical of very well made
pure class A1 PP triode
with 2 x KT88 in triode and with 20dB global NFB.
However, I have to say, "GEE, what a huge amount of NFB!"
Is it all really needed? Could it ever really be used? Well, in
fact, possibly because I
have NOT BUILT THIS AMP, but I can see already that there could
at both LF and HF just outside the audio band. The fact is that
the Hammond 1640SEA
does not have extremely low leakage inductance. Its barely low
enough for general
conventional use let alone for the "sophisticated" schematic I
have proposed here.
But if anyone made an OPT with twice the amount of interleaving
used by Hammond,
they may surprised by what might be done. In general, I have
found the 16xx series
SE Hammond OPTs to be very useable and good sounding.
In any amp, as the total amount of open loop gain without any FB
increased, any application
of NFB tries to extend over a wider range of F thus extending
bandwidth beyond the
open loop bandwidth. For example, if OL BW is from 30Hz to
20kHz, 10dB applied GNFB might
extend BW from 10Hz to 40kHz. 30dB GNFB may increase BW to be
2Hz to 150kHz.
But the phase shift of the open loop amp at LF and HF will cause
the applied FB to become
positive and hence cause oscillations unless gain shelving
networks are used to reduce the
phase shift and reduce the open loop gain outside the audio
band. There is no need at all to
have a high amount of NFB applied outside the audio band.
In the Fig 6 amp, probable open loop F1 pole = 40Hz and F2 pole
= 5kHz to get LF and HF
This means that the high amount of NFB only applies to the band
of 40Hz to 5kHz.
However, perhaps music is better and amp more stable to have
less open loop gain and
less total NFB, with slightly more THD and slightly lower
damping factor and open loop
F1 and F2 further apart at 15hz to 20kHz, resulting in final
bandwidth of 7Hz to 65kHz.
There is a simple answer, just
leave out the screen NFB and use the EL34 strapped
as a triode. EL34 triode grid signal will be 4.5Vrms. 12AU7 can
have a higher FB signal
applied to its cathode = 1.5Vrms. Total Va to Vk = 6Vrms, so
Vg-k = 0.4Vrms, so input
signal to 12AU7 grids = 1.9Vrms. This seems high, but is OK, and
the amount of GNFB
= 13.5dB, and the amp's THD at 25Watts = 0.4%, quite good.
Now take note that in all my amps I have supplied to customers,
I have provided
protection circuits to prevent damage to OPTs and other parts if
one or more output
tubes decides to conduct far too much Idc because for one reason
the grid bias voltage ceases to control the Ia flow. Hence the
note on the drawing
about tp1, tp2, tp3. A KT88 which becomes a short circuit could
damage the cathode
CCS mosfet. I show no cathode fuse because before it would blow,
may fail with excessive Vd-s and enough current to fuse the
There is a zener diode in the mosfet, but it could fry
OK, now you have seen the "traitors way to use Squalid State
devices" in a tube amp.
How could I ? Didn't I know they don't belong ?. Well, OK, I
hear the complaints,
but constant current sources or current sinks using SS ARE OK,
because they are
so good at providing an extremely high impedance source of
current and as such
cannot have any effect on the signal. The SS devices are
friendly slaves, totally
under control of tubes, and they allow tubes to work better than
they other wise might.
Now, to appease the "Society Against Solid State", I have
another similar schematic
below which is similar to Fig.
3 x KT88 may be used with Hammond 1640SEA but with a single
choke in cathode
circuit as seen in Fig 7 below.....
Fig 7 is the same as Fig 6 but has SS current sinks replaced
with a choke.
Ah, so simple! - until you start thinking about a choke, and go
shopping for it!
If you build this type of circuit using an
then all I said above about how it works applies.
The Fig 7 amp like those above have a high total amount of NFB.
But to prevent oscillations with a high amount of NFB you need
to be able to
Nyquist and Bode graphs and theory intuitively. Its too
difficult for me to
write a book about it right now and have you read 2 pages to
So, do your own Googling with Bode and Nyquist, and lose a week
of your life
trying to understand WTF it means.
Basically, as one increases the amount of NFB and or the amount
of open loop,
gain, ie, the voltage gain from input to output without any NFB
applied, then the
amp becomes more prone to oscillations due to the phase shift
reactive circuit elements L and C reacting with R. For most tube
amps there is
ONE F where there is no phase shift and it usually is between
200Hz and 2kHz.
But below 15Hz and above 15kHz, and where the open loop gain has
attenuated by R&C Miller effects, C&R couplings, OPT
primary L, leakage
inductances, shunt C, etc, and where phase shift reaches 180
degrees and where
gain exceeds 1.0, or "unity", then the amp will oscillate.
The only way to prevent this is to use Zobel networks at output
and perhaps in
output anode circuits and at V1 output.
In Fig 7, the R&C values affecting stability ARE A GUIDE
ONLY, and YOU have
to figure out the best values for unconditional stability.
The critical R&C parts are R12&C8, R16&C9,
C6&R13, C19&R35, C21&R37.
If you find you just cannot stop oscillations, then abolish the
screen FB to EL34.
Connect bottom of C12 to 0V instead of to the FB from OPT.
Reduce value of R16 to 1k0 to increase GNFB.
But you will still have to optimize the bandwidth with a pure R
get a good square wave without severe ringing or HF oscillations
pure C load between 4uF and 0.047uF.
Cathode FB with a custom
It is time to mention my favorite output stage configuration for
both SE and PP tube amps.
Such stages have 2 primary windings and the usual secondary. One
of the primary
windings has up to 1/3 of the tuns of the other and is called a
cathode feedback winding,
or tertiary winding.
Conventional local CFB in SE output stage may be used if you can
find an OPT which
has a separate cathode feedback winding but really it is just part of the
winding because the tube signal current flows in both windings. But the CFB
is usually at an earthy Vdc potential, while anode winding is at
the B+ Vdc.
Fig 8 has two examples of CFB
in output stages.
Fig 8 Left side.
The total signal voltage across both primary windings = 184V +
46V = 230Vrms.
20% of the total turns are in the CFB winding and 80% in the
I show the formula for working out the effective UL % as
UL% = 100% x ( V ULtap + Vk ) / ( Va + Vk ) where the signal
Vrms are measured
between each of the screen, anode and cathode terminals to 0V.
In this case the V UL = 0V because there is no UL tap. But the
KT88 is still
operating as though there is a UL tap because a signal of 46Vrms
screen and cathode. Pure beam tetrode or pentode with CFB is
bypassing the screen 100uF to the cathode so there is no signal
screen and cathode. It is usually found that the operation of
pure beam or pentode
with CFB merely reduces the complex THD spectra of the pure beam
The use of an effective UL tap by means of bypassing the screen
to 0V changes
the THD spectra towards triode with less odd H. Usually, where
the CFB % of
total primary turns is close to 10%, the effective Ra of most
beam tetrodes and
pentodes to be about equal to the triode Ra. When the CFB % is
10%, the Ra can become considerably lower than triode Ra.
Optimum CFB %
will be about 20% but can be from 10% to 40%.
The higher the CFB%, the higher the drive voltage needed so the
have low THD.
Usually, the driver is a trioded EL34 or EL84, but could be 45
or 3A3, although
real triodes have very low gain so a high gain input triode is
needed. So in my
view the EL34 or EL84 are winners. Inevitably, driver triodes
2H. If the driver has to make 75Vrms, expect 0.7% 2H, and the
may make 1.5%. The 2H generated will cancel, and 2H total at
1.5% - 0.7% = 0.8%.
However, the 2H of all such CFB use with KT88, EL34 varies with
load, and there
is an RLa value just above the RLa value for maximum possible Po
where 2H = 0%.
Below this RLa value the 2H cancels, above the RLa value it
adds, because the
relative phase of the 2H of the output tube is the opposite
It is more fully explained at my pages on the SE35 amp.
Fig 8 Right side.
The operation of the right side has effective UL % = 40% because
there is a UL tap
at 20% of total primary turns and there is 20% CFB.
With class A operation there could be a UL tap at 40% of total
turns which would
increase the effective UL total to 60%. Going beyond this % is
pointless because it
restricts the Po available because operation becomes too
much like a triode
where the Ea negative swing is restricted by grid current onset.
But at over 40% effective UL, the odd H are very much reduced
leaving triode like
THD which is mainly 2H, and the phenomena of having 0% 2H at a
diminishes and you get good cancelling of 2H produced by the
For SE class A1 and with most beam tetrodes and pentodes the
about 20% CFB and UL seems to give extraordinarily good sound,
so I have
"Shunt feed", aka "parafeed"
Shunt feed output stages have seldom ever been used for hi-fi
need a large air gapped choke with high Idc flow, high
inductance and not likely to
saturate with a high anode signal voltage at above 14Hz. This
choke provides a high
impedance source of Idc to the tube while not consuming any
audio power produced
by the tube. The size and its weight may be larger than the OPT
The audio signal power of the tube is conveyed to a primary
winding on an ungapped
OPT via a coupling cap. The OPT primary may be connected to 0V
at one end,
and the cap has a large Vdc voltage across it. The OPT can be a
normal PP OPT
which is very easy to source, and its Lp inductance is usually
far more than is actually
needed, and it usually does not saturate above 20Hz at full Po.
The coupling cap
capacitance value must be high enough to have a resonance with
OPT LP at below
3Hz. So if Lp = 30H at low signal levels the C should be 100uF.
One might use a
number of C in series with resistance dividers to ensure equal
Vdc is across each
cap and Vdc never exceeds 2/3 the Vdc rating for the cap.
Electrolytic caps are
needed, but each must be bypassed with 1uF plastic film caps.
The Shunt Feed advantages are
The OPT has no huge Vdc
potential between primary and earthy secondary or
0V or chassis.
The OPT may have no air gap.
Laminations may be maximally interleaved.
OPTs meant for PP amps may be used.
For the maximum Po, the OPT may be smaller than if the
OPT was a conventional
air gapped OPT with Idc flow.
The Shunt Feed disadvantages are :-
A large choke or 2 or 3 series
chokes which may have the greater size and weight as
the OPT must be used between anode and B+ to provide a high impedance feed of
Idc to anode.
Capacitor coupling from anode to OPT must be used which
introduces yet another
time constant filter behaviour which can affect the LF stability
of the amp when NFB
is used. GNFB
application with a choke feed amp may be more difficult because
unconditional stability must be assured. The amount of GNFB is
limited by the number
of C&R and C&L couplings and L&R shunts. But with
triode output tubes 12dB GNFB
with no output load is usually enough, and possible, when LF oscillations are most likely.
More careful arrangement of open loop gain shelving R&C networks
But for HF stability, there is no extra stability problem
compared to using a
gapped OPT and conventional GNFB arrangement.
The capacitors from anode to
OPT must be chosen carefully and used with respect to
their voltage ratings.
Fig 9 shows how choke feed allows the use of a PP OPT without DC
the primary winding.
I show the anode choke L1 = 55H at 80mAdc, and estimate Rw at
cannot purchase a Hammond 55H choke equal to what I say is
Hammond have have 193C 20H with Rw 180r so 3 in series are
needed for 60H.
But then Rw total = 540r giving 46Vdc and B+ must be raised from
+430V to +460Vdc.
So before you copy what I show, be sure you know ALL about what
you are doing!
The screen is cap coupled with 100uF to the CT of OPT primary
giving 50% UL.
The screen requires low Idc
of 4mAdc choke feed through L1, perhaps a Hammond
155C, with Rw = 2k7, L = 60H.
I estimate 50H is plenty, and to get the Eg2 nearly equal to Ea,
and to prevent fusing
the L1 choke if screen shorts to cathode or 0V the series R =
1k0, 0.25W rated.
If 300Vdc appears across 1k0, it fuses quickly, and a new R
All electrolytic caps used should be rated for 350Vdc, and each
bypassed with 2uF,
plastic film types rated for 630V.
The C value for coupling anode to OPT seems quite high, 110uF in
The LF pole formed by 110uF and anode load 4k2 is 0.34Hz, but
more important is
the pole of HPF formed by the 110uF and OPT primary L which may
at low levels of signal typically used for listening. This F
pole would be 2.14Hz,
and the peak in response is prevented by the very low impedance
and OPT at 2.14Hz. Therefore enough global GNFB should be able
to be applied
while maintaining unconditional LF stability with R&C
critical damping networks
in input - driver stages.
Triode Choke feed.
The triode use of KT88 can have B+ at a higher voltage for best
a 300B could also be used instead of the KT88, although the B+
has to be +40Vdc
higher because the Ek bias will be about +88Vdc. The triode use
does not need
any screen choke, but anode choke needs to have high L value as
All the same comments made about SEUL apply about the L1 and
Triode is easier because the screen is simply strapped to anode
via its stopper
resistance of 220r.
845 Choke feed.
An 845 may happily work with an easy to buy Hammond 1650P OPT
60W, but used to make about 21 Watts. The load match is 6k6 :
To make an air gapped OPT for the Fig 10 output stage with an
845 is extremely
difficult for 99% of DIYers. Making their own chokes would also
be very difficult.
But they could purchase 1650P OPT and three Hammond 193C chokes, and the
required capacitors. Possibly MUCH better chokes and PP
OPTs could be found
than made by Hammond, but appraisals of whatever brands are used
full understanding of how the basic item properties affect the
I won't suggest how an air gapped OPT or choke may be designed
but the method and examples can be found in my other OPT design
I must mention the analysis
behind Choke feed.
A choke is
defined as a coil of enameled wire. Its basic properties are
an amount of pure inductance in Henrys in series with resistance
= winding wire ohms.
There is also capacitance between turns resulting in a summed
effect of an amount C
shunting the L. So all chokes have a parallel resonance between
the L and the C.
For a single choke of 55H with an iron core, shunting C might be
The reactance XL of a 55H choke at 10Hz = 3,454r, and far less
than the C.
The XL rises with F to a maximum of perhaps 500k ohms at the
Fo at 1,240Hz. Above Fo, the XL reduces because the XC declines
with rising F and
= 3,454 ohms at 153kHz. To extend the high XL the choke can be
wound on a bobbin
divided into say 3 sections physically 2mm apart so that the
100pF shunt C of each
is in series with the other sections thus reducing C total to
In practice, the C has little effect when the choke is in the
anode circuit of a tube
Chokes oppose the flow of AC because the voltage across the
choke sets up a
magnetic field which acts to oppose the flow. Therefore low
current flows in the
choke across the audio band. The small amount of audio power
that is lost
= Iac squared x Rw, and is usually negligible. The same applies
to the inductance
of any OPT where the power lost as heat in the OPT = Iac squared
The OPT and choke Rw losses are greatest at bass frequencies
where Iac becomes
highest and XL is lowest.
The shunt feed choke may be two or 3 chokes in series to make up
the wanted total
choke inductance as I have shown in Fig 10.
The total L is simply the sum of the individual L values. The L
values don't have to
be equal, one could have a 40H choke plus 20H choke to make 60H.
But they MUST not saturate with the presence of large signal
voltages at LF
and combined with high Idc flows.
For Shunt feed SE amps, one aims to ensure core saturation of
choke or OPT does not
occur at F higher than 20Hz and with a signal voltage at the
maximum Po level for the RLa
value which gives the highest Po at the clipping level.
For an 845, one might use RLa = 6k6, and get at least 21Watts so
Va = 372Vrms.
The 1650P OPT is rated for 60W to 6k6 and OK for 629Va-a and
saturation could be
at 30Hz. ( I am not exactly sure. ) But saturation F is a
voltage dependent phenomena
and occurs independently to loading and currents. So at 372vrms,
expect the Hammond
OPT to saturate at 18Hz. The maximum primary inductance at high
Va-a is probably more
than 300H at say 1Tesla, typical with GOSS core material with
The choke has to cope with 372Vrms at 20Hz without total Bac and
Bdc summing to
more than say 1.2Tesla which is a maximum for medium grade iron.
Top grade GOSS
lams or C-cores may take only 1.4Tesla before onset of
core saturation and poor old iron
from 1950 might take only 1.0Tesla. If you make the3 choke,
assume the maximum iron µ
is below 2,000, and that it saturates at 1.0Tesla. So one may
find one has to wind TWO
chokes and connect them in series. One may buy chokes, but
whatever is manufactured
or bought MUST satisfy the engineering design requirements, and
whatever is used must
be tested carefully.
The total L shunting RLa is the choke L plus OPT Lp in parallel.
In all shunt feed SE amps,
the air gapped choke will have much less L than the cap coupled
The total L should have reactance = RLa at 20Hz or lower F.
Minimum Choke L value = RLa-a / ( 20 x 2pye ).
For RLa = 6,600r, L should be 6,600 / ( 20 x 6.28 ) = 52.5H.
During the amp's life it will be used at less than 1 Watt, so
the Vac across the L will be
less than 1/5 the clipping level. The inductance of iron wound
coils varies with the
applied voltage, ie, the lower the Bac, the lower the
permeability µ. Air gapped chokes and
air gapped OPTs have the least variation and reduction of L at
low signal levels. So
we don't need to aim to make the calculated choke L higher to
compensate the drop in µ
at low Vac levels. Most variation of L occurs with non
gapped cores for PP amps where
inductance at 0.05Watts, 18Vrms across 6k6 load, may be 1/6 of
the 20Watt Lp level,
and so may be 50H. This L in in parallel with choke L and if the
choke L = 50H then
the total L may drop to below 20H at very low levels. This WILL
NOT cause any change
to the F response or bass performance because the tube Ra also
shunts the L and the RLa,
and the 845 Ra = 2k2 plus 6k6 RLa makes the R shunting L = 1k7,
so F1 pole with 20H
is at 13.5Hz. The 845 ( or any other tube ) will easily drive a
low inductance load at very
low signal levels.
However, most people will wish to use NFB and the reduction of L
at low levels causes the
F1 pole to rise at low levels despite the Ra and RLa shunting
the L and the additional phase
may cause LF oscillations. Many old mass produced tube amps
require a speaker to be
connected while the amp is turned on lest they begin to
oscillate at some low F below
10Hz. I have often encountered such amps and the oscillations
can be violent enough
to cause core saturation and heavy damaging tube currents, or
else the amp oscillation
is limited to a low level because inductance rises with signal
voltage so F1 moves down
and so does circuit gain so the amp in a state of equilibrium as
a phase shift oscillator.
Connecting a speaker puts more R across L and reduces F1 and the
oscillating. Such amps have been designed by accountants. But
now you see the need
for LF gain shelving R&C networks to prevent the
oscillations at LF.
Above I said total L minimum = 52.5H. If the OPT Lp was in fact
say 300H at 21Watts,
then we should prefer to ensure total XL = RLa-a at no higher
Therefore the choke L should be 63.5H, so when in parallel with
300H the total = 52.5H.
Basically, we want all the choke L we can muster!
Design of a single choke could be....
Ia = 83mAdc, so chose wire size so current density = 2A/sq.mm
for where Idc = 3 x Iadc.
For 249mAdc, wire size = 0.398mm Cu dia. Use high quality
magnetic winding wire, 0.4mm Cu dia, overall size including
enamel = 0.47mm dia.
Winding window available = 20mm x 72mm = 1,440 sq.mm so random
possible = 1,440 / ( 0.47 x 0.47 ) = 6,518 turns. Expect to get
6,000 turns on.
Experience tells me that probably this will give enough L at
83mAdc, and with the correct
air gap. But YOU need to read my pages on choke design to verify
that the choke design
is correct. The L is adjusted for a maximum by adjusting the air
gap size with 83mAdc
present plus a 50Vrms Va signal at 50Hz, without any load
connected at OPT.
Perhaps you may find choke L is higher than you wanted, which is
great news, and bass
will be fabulous.
To make at least 2 chokes, one for each stereo channel, you
would need about 2Kg of
new copper winding wire, 2mm fiber-board for making bobbins, and
laminations. These may be taken from a large power transformer
with a fused winding.
I have often used old laminations from defunct PTs for chokes.
To easily take the laminations apart, the transformer should be
placed in a small wood
fire for just long enough to make the lams appear dull red. The
varnish and all plastics
will be vaporized and burnt. The burnt out core is left to cool
slowly and will easily
fall part when the bolts an wire are cut off. The firing is a
messy process best done
late on winter nights if you have a fireplace. The heat may
improve the magnetic
properties, but won't worsen magnetic properties.
There are other methods
of avoiding the high +Vdc potential between an OPT primary
and the earthy secondary. The first uses a "floating" B+ Vdc
that say the earthy
rail of the supply connects to OPT primary at the normal B+
connection, while the other
end connects to 0V. The positive rail of B+ supply connects to
output tube anode/s.
This means the HT winding on PT and its diodes and filter caps
and chokes all are
at an elevated B+ potential and all these items carry the anode
signal. There is some
capacitance between all this hardware and 0V and chassis. There
between secondary HT
winding and other PT windings such as mains windings which
may introduce noise to
the anode audio path.
Therefore the PT should have extremely good insulation between
its HT winding and all
others especially if the B+ supply was 1,200Vdc for 845 or 211
tubes. If the maximum
possible peak signal voltage swing at any given instant was
+1,100V, and the adjacent
mains voltage was -340Vpk, and the B+ was +1200V, then one could
between windings. Hence the need for good insulation. To
minimize HF diode switching
noise signal getting into the relatively high impedance anode
circuit the PT should have
an an electrostatic
shield. An ES is usually one layer of wire turns with one end
to 0V and other end left open. Many old radio sets had such a
shield on their PT.
The amp heater windings can function as a shield if one whole
layer of thick wire is
designed to be devoted to heaters which have a CT connected to
So thus the PT can become a custom wound item.
Building a PSU that is
alternative to the normal arrangement of grounded B+ supply is
more difficult and expensive, and cannot improve sound quality.
Fig 11 shows a KT88 operating in SEUL with a floating B+ supply,
but also has
a choke feed. It is yet another way to avoid having a special
expensive OPT with
air gapped core.
I bet nobody else has ever used the schematic I have here.
Certainly I have not
ever seen a mass produced sample. Manufacturers always will try
excessive iron wound items and any extra components that could
be avoided by
designing a good OPT with Idc flow and air gap. Often their
attempts remind us
that an accountant designed the amp, not an engineer. Keen
DIYers need not
heed any accountant's advice and they may be free to design
according to basic
principles and be creative in the process.
The audio is amplified in the same way as for any standard SEUL
But the choke feed is arranged differently to standard choke
The B+ supply is floating, ie, not grounded anywhere, and I show
the PT with its
HT secondary, diodes and 470uF caps and filter choke all
operating as a +433Vdc battery
between a KT88 anode and a choke at 0Vdc potential. The Idc path
is from positive rail
of B+ supply to anode then down to cathode, through R&C
cathode biasing network, then
to 0V rail then through L1 "choke feed". The B+ PSU for +433Vdc
has the anode signal
present at all parts, yet the floating PSU acts entirely without
any interference from the
audio signals, and does not impart any noise into the audio
There is 0.85Vrms of 100Hz ripple across reservoir caps, and
1.1mV across the
pair of caps for +433Vdc.
The feature of noiseless floating B+ PSU is made possible with
the use of an electrostatic
shield on PT between the B+ HT winding and other nearby windings
with diode switching
Old amps and radios had a shield made with one layer of thin
wire between Mains input
primary and HT winding. One end of the layer was taken to the
amp or radio chassis,
with other end left open. But the tube heater winding of 6.3V
can be made to occupy one
layer and have its CT taken to 0V rail and it acts as an ES
between mains and floating
B+ HT winding. But an ES should be wound over this floating
winding if additional B+
windings are which have diode rectifiers. An ES need not be a
layer of wire, but can be
one turn of thin copper or brass foil of say 0.1mm thick and
overlapped 10mm, but
prevented from being a shorted turn because insulation is
between the overlapped
turn ends. A wire lead is brought out to 0V. Therefore the only
energy transferred to
the floating HT winding is magnetic. I've shown a voltage
doubler type of B+ supply
with CLC filtering and 1N5408 Si diodes, quite good enough.
The tube operates with Ea = 375Vdc, and with +418Vdc at its
anode and with +43Vdc
normal cathode biasing and grid at 0Vdc potential. The anode
signal is transferred to
the top of choke L1 55H which has one end connected to 0V.
Notice that the Idc
through L1 produces a NEGATIVE Vdc across the choke. The higher
the choke Rw,
the greater this -Vdc will become, and the higher the B+ supply
voltage will have to
be to ensure the wanted Ea is produced. So low resistance
choke is wanted.
The L1 choke of 55H does not significantly load the KT88. 55H at
XL = 345k plus some shunt C of say 300pF = 530k. Now the
proposed ideal RLa for
maximum Po = 4k2, so XL = RLa at 12.15Hz, well below 20Hz, so at
full 12.6Watt Vo
level the KT88 does not put much signal current through L1 above
Fig 11 shows details of L1 and L3 chokes. L1 could possibly be 3
x Hammond 193C
20H chokes in series but then Rw = 540r total and the L1 Vdc = -
46Vdc, so B+
must be raised from +433Vdc to +464Vdc, which means +496Vdc must
by HT winding so the highest voltage may not be high enough at
The 193C have maximum Idc rating = 100mA, which is a bit low,
and the DIYer can
make a MUCH better choke at home.
I hope everyone fully understands that if you change one thing
properties of one component, it can seriously affect more than
one other thing and
perhaps ruin the optimum operation of the circuit. SE amps with low max Po
need all the optimization possible so that a wide range of loads
can be used.
The SEUL schematic has the anode signal appearing across L1, and
to get the
wanted 12.6Watts of audio power there must be an R load across
L1, and it is the
OPT which is cap coupled to the L choke with 1,000uF. The
1,000uF prevents any Idc
flow in the OPT primary which always remains at 0Vdc potential.
Now the top of choke
L1 is at -15Vdc and the 1,000uF cap may then be a low voltage
type rated for say 35Vdc
and best are those with high ripple current rating normally used
in PSU for SS amp circuits.
The Vdc available will adequately polarize the electrolytic to
give linear operation. A 2uF
plastic film cap should be strapped across the el-cap to shunt
its ESR, and make the cap
work as a pure cap to 10MHz.
The OPT I recommend is Hammond PP type 1615A, rated 5k0 : 4r, 8r, 16r.
The primary CT is used for the UL screen connection. But the
screen must have the same
+Vdc voltage as the anode, so it is coupled to the OPT CT with
100uF el-cap, also
bypassed with say 1uF/630V plastic film. To get the screen to be
fed with Vdc = anode Vdc,
I suggest the L2 choke, over 50H. It can be a Hammond 155C,
giving 60H, but its Rw is
high at 2k7, and with screen "stopper" resistor 1k0, and Ig2 =
5mAdc, the screen Vdc is
18.5Vdc lower than anode Vdc. Best SEUL operation occurs with
Eg2 = Ea when nearly
all power beam tetrodes and pentodes need Eg2 = Ea lest the
maximum Po be restricted
similarly to triode operation. Ea to Eg2 difference should be
limited to less than 5% of Ea.
There is no doubt that a keen DIYer could make a slightly larger
choke than the Hammond
155C and with MUCH lower Rw using thicker winding wire and a
Such a choke is less prone to fusing open if a screen becomes
shorted to cathode,
or the coupling el-cap to OPT becomes shorted.
This L2 choke also has 1/2 the anode signal across it, so it
must not saturate with
excessive total Bac + Bdc.
Fig 12 gives the choke feed arrangement for floating B+ and
It is very similar to Fig 10 for SEUL. The KT88 screen is
connected to anode
with 220r stopper, so that the screen is being fed ALL the anode
The screen g2 acts as an additional control grid in the same way
in a "real" triode acts to partially control Ia in conjunction
with the g1
control grid. Triode operation gives the maximum possible amount
NFB introduced into the tube, so the voltage gain is lower than
The SE triode operation is obviously simpler than SEUL, and the
output power is not much less than SEUL. But notice that to get
near the SEUL Po, the triode Ea and Ek must be higher with Ia
for the Pda to be 30Watts, same as for the SEUL use.
If the KT88 strapped as a triode was used with exactly the same
Ea and Ek as
shown for SEUL in Fig 10, the Pda + Pg2 would still need to be
kept at 30Watts.
The Ia+Ig2 = 85mAdc, so for maximum Po the RLa = ( 375 / 0.085 )
- ( 2 x 1,100 )
= 2,211 ohms. This RLa is only 2 x Ra of the triode. THD is
higher than SEUL.
Max Po at anode = 8Watts. Damping factor = RLa / Ra = 2k2
/ 1k1 = 2 which and
no better than the SEUL with 4k2 load and 50% UL tap.
Then the OPT which was chosen for UL has nominal ZR = 5k0 : 4,
and for RLa = 2k2 triode anode loading, required secondary loads
= 1r8, 3r5, 7r1.
But the winding losses will double to become 20%, so therefore
the max Po with
triode at the secondary becomes about 6.4Watts only.
Therefore I hope everyone can see that to get the KT in triode
to work nearly
as well as SEUL, you MUST use a higher Ea and lower Ia to suit
the load which
is nearly 4 x Ra, and OK for good triode operation.
Now we might consider the KT88 in SEUL mode with Ea +440V and Ia
+ Ig2 69mA.
The Ia will be 64mA, and RLa for max Po = 0.9 x 440 / 0.064 =
Max Po at anode = 12.7Watts. The OPT secondary loads needed to
give RLa = 6k2
are 5r0, 10r0, 20r0. A speaker of "8r" could be used on the 8r
outlet, but the primary
loads becomes 5k0, and Po max = 10Watts. But an 8r speaker may
dip to 5r0 so it
is best used on the 4r outlet and the tube will produce highest
power where the Z dip
occurs which is OK. But the use of a "4r" speaker with a dip to
3r will reduce RLa to
3k7, and Po max = 7.5Watts with high THD.
However, all this isn't as bad as using the KT88 in triode with
Ea = 375V.
SE amps can only give maximum Po at ONE load value. Suppose the
12.7W max is 3r5, when a "4r" speaker is used on the 4r outlet.
Then if speaker Z
dips to 2r0, Po max about 6.7Watts, and if Z rises to 8r0, Po
max = 6.6Watts.
So although a "12Watt" amp can make 12Watts, the inevitable
variation in speaker
Z across its frequency band and the low sensitivity of modern
speakers will reduce
real capability of the amp to 6 Watts. Hence high sensitive
speakers should be used.
However, with say 3 or 4 x KT88, the usable Po range becomes say
and then the load matching becomes far less critical and the amp
well with any speaker.
Back to Education and DIY
Back to Index page