Graph 1 shows ESL57 impedance vs frequency and is fairly accurate.
varies so much the power needed varies with
The Quad-II amp can develop enough output voltage at low F to get sufficient class A1
power for F between 40Hz and 100Hz and clean class AB1 power for 100Hz to 1kHz.
Above 1kHz power needed reduces while at the same time the speaker Z reduces so
there is full range operation without much overloading or clipping most of the time.
Music energy above 2kHz rapidly declines.
Quad ESL57 have minimum Z 1.8r at 18kHz and amp can make only 5Watts with load
of 1.8 ohms. But Quad-II amp copes with making a low amount of Po for HF while
making most of its power at lower F where Z is above 8r0. Therefore Quad-II is fairly
well matched to ESL57 in the main power band. The more anyone thinks about ESL57
and tube amps, the more they should realize ESL57 are easy speakers to drive, and that
a humble tube amp can power them very well, ( providing teenagers are not allowed near
the volume or tone controls ).
2007, I was asked to examine a pair of Quad-II amps. One overheated
during summer weather. In both amps I removed the single 180r Rk and the shorted
Ck 25uF which troubled one amp. I installed bias networks to each KT66 cathodes,
470r plus 1,000uF/63V between cathodes and ends of CFB windings. The CFB winding
CT was taken to 0V. The amp then ran with better regulated Ek for each KT66. I found
no other faults.
But one amp
failed again after the initial repairs. One had burned one 470r open
one of the 63V rated 1,000uF caps had shorted, and after only an hour powering the
owner's pair of ESL57 speakers. The KT66 survived the experience because the owner
turned off the amp soon enough. I examined his ESL57 speakers and found one with
arcing mid-treble panel, a common fault in ancient ESL57, and expensive to repair.
The owner had bought the ESL57 on Ebay from UK, and Ebay deals are always risky.
The panel arced when signals went above about 2Vac. Arcing of a panel means the amp
has a short circuit for a load during the arcing, and this causes excessive signal currents
and dc currents in the amp. One KT66 went into "thermal runaway." The 1.0 amp mains
fuse did not blow. The the mains transformer became so hot it expelled some of its
internal tar based potting compound. What a mess! This clearly illustrates the messy
way in which old tube amps can fail because of faulty 50+ year old speakers, and the
absence of any active protection circuit. I repaired the amps a second time, but fitted
active protection circuits. I think the amp owner bought a new pair of dynamic speakers.
Quad-II have been said to offer fine class A performance, and the amount of class A
depends on the idle Pda of the tube, and with KT66, up about 40% of the Idle Pda
can be converted to class A1 audio power. Each KT66 generates about 24Watts of
at idle, even without any audio signal, so there's a total "Pda" of 48Watts, and about
45Watts is anode heat, with the rest as screen heat. So with the ideal load value, the
tubes can make 18Watts of class A and because OPT winding losses are about 10%,
you get a possible maximum of 17Watts class A1. However, the speaker loads for
maximum class A need to be twice the loads for the nominal OPT strapping of OPT.
For example, if the OPT is strapped for 16r, the load for max class A = 23r, and for
8r0 strapping the load should be 16r. So where someone uses a pair of old 16r
Tannoy speakers with higher sensitivity than ESL57, Quad-II is happier and performs
better with 8r0 strapping.
Graph 2 shows the maximum power levels where THD < 1.5% and available with the two advertised
methods of OPT strappings for either 9r0 or 16r0, and the never advertised strapping for 4r0.
Fig 1. The original Quad II
monobloc amplifier schematic :-
Part numbers are the same as the original schematic.
I scanned a
good 50 year old paper copy of the schematic and here is the re-drawn
version I hope everyone finds easier to read.
Two OPT strappings are touted for Quad-II.
One way = "16
ohms", other is for "8 ohms."
only speakers over 8r should be used with the 8r0
strapping, and only
speakers over above 12r should have OPT strapped for "16r0", including ESL57.
For all other speakers only the 8r0 strapping should be used. ESL57 will work with 8r0
strapping at slightly lower maximum Po.
Note. The nominal tube loading is 3,888 ohms anode to anode, ie, across the whole
of all OPT primary anode and cathode windings. This is with 16r0 load at the 16r0 strapping.
or 9r0 ohms at the "8r0" strapping.
There is no
recommended strapping for a "4r0" match which would best suit
of speakers made after 1980 when speaker sensitivity was reduced and power handling
increased because of cheaper power from SS amps. Although I might upset many owners
of Deare Olde Speakers made before 1980, their F response and general sound quality
was often quite poor. Modern speakers often have Z between 3r0 and 10r0, with dips in
Z to 3r0 within main power band of 50Hz to 500Hz.
An "illegal" load match to 4 ohms is possible if the wire from speaker terminal to point T
on the OPT is moved to point Q. Point Q uses only 1/2 the available secondary turns and
its use with 4r0 means winding losses are high. Peter Walker did not intend to confuse ppl
more than they were by OPT strapping. A few Quad-II have been strapped for 16r, and
ended up on my bench after someone tried to use a modern insensitive 4r0 speaker at loud
levels. People forget the strapping setting, and they have ZERO ideas about anything
When using output from Point Q, the existing links between T, S, R, and Q may all be
ignored because the two windings between T and Q cannot be used.
I think Peter Walker made a mistake by not having one more turret terminal on the OPT to
allow all the secondary turns to be used for a 4r0 load matching, and thus be able to to keep
the winding losses as low as for when 16r0 speakers are used with strapping 16r0.
high winding losses with the "illegal" 4r0 link of speaker to
point Q, the power
available is not much lower than for other load matching. Graph 2 shows the measured
power and it speaks for itself.
overload behaviour is fairly benign. Most PP output stages with
will suffer increasing Ek when driven hard in class AB. This means the grid bias is increased
and the tubes become over biased and THD can increase tenfold. But in Quad-II, I found that
it was difficult to cause more than +10% rise in Ek with any load up to clipping where grid
current in KT66 had just begun. The B+ did not sag more than -10% when clipping was reached.
The EF86 clip when grid current begins because they cannot drive a low Z load. I conclude
Quad-II has fairly good inherent ability to withstand BRIEFLY excessive signal levels.
Most output tubes withstand brief excessive saturation where instantaneous Pda or Pdg2
exceed ratings temporarily. But if Vdc and Idc conditions change for long enough, output tubes
will overheat badly and a tube or two is doomed unless the amp is turned off.
To get lower
winding losses requires different OPTs. I've looked everywhere and found
there are no replacement OPTs for Quad-II which offer better better performance which
will fit inside the original steel sheet can.
I now have a pair of custom wound OPTs made with cores much larger than those used
by Quad, and which will be used in a pair of Quad-II with a totally re-engineered schematic
with KT88, triode input stages, and new layout for tubes without GZ32 rectifier or choke.
If the original Quad-II OPT is retained, better technical performance and sound is
if the original circuit is upgraded with modern R&C parts unavailable in 1955 and including
high capacitance electrolytic capacitors. There is no need for GZ32 or GZ34 which are best
replaced with silicon diodes. The use of a couple of LEDS and a couple of small bjts can
now be used for protection circuits and bias balance indication to make sure an owner
knows how his amp is going, and if there is a faulty output tube. This is prudent in an age
where modern people are just not used to the unexpected and perhaps smoky failure of
the primitive amplifiers of the 1950s, and there is now no Quad Company Support guy
you can depend on for service or new Quad parts.
quiescent bias anode currents of the output tubes change as
tubes age. The two
currents easily so different because there is only one shared "cathode bias" network
of R12 180r and C5 25uF. And with aging, the tube grids begin conducting small but
unwanted grid currents even at idle so the Eg1 may rise to a positive Vdc above the
bias supply point at top of R10, 100r. The Vdc measured across R10 should be about
+0.23Vdc. The chosen value of grid bias R7, R9, of 680k is much too high. After 50
years, typical R values go to 750k. The high values were chosen to allow the weak
EF86 to operate without reducing the voltage gain by having RLa too low. But the
+Vdc which appears across the 680k in an aging KT66 causes the idle current to
go higher which raises tube temperature which causes even more +Vdc across 680k
and and more heating. This is an unwanted positive feedback mechanism.
The Vdc across the 680k measured normally should be no more than +1V, so
between KT66 grids and 0V, you don't want to see more than +1.25Vdc. I have seen
KT66 at near the end of their life with +9V at the grid at idle and with 90mA of anode
current with slightly red hot anodes. This is disastrous for the music, and such a tube
continues to overheat insidiously before finally melting down internally, and perhaps
terminally damaging a power and / or an output transformer. If ONE KT66 begins to
conduct too much Ia, the Ek rises with high current in R12 180r. This rise in Ek tends
to turn off the Iadc in other KT66.
The output tubes rarely ever age at the same speed. So while 90mA may be flowing
in one tube, there may be 40 mA in the other and there is a net 50mAdc flow across
the primary which can magnetically saturate the OPT core to cause bad distortion
because the OPT core has no air gap and was designed for well balanced and equal
Iadc = 70mA in each output tube. The effects just described can be worse if the old
Hunts 0.1uF coupling caps to EF86 anodes have become leaky, further increasing the
positive grid voltage. I recently found two such 0.1uF caps had become 400k resistors.
The KT66 had high +Vdc at each grid, saturating both KT66 to have about 500mA flow
in R12 180r, so it fused open fairly soon.
EF86 input tubes are set up in the
original amps in what is called
a "floating paraphase
phase inverter". It means a fraction of the output from V1 anode is applied to V2 grid to
achieve two equal amplitude drive signals to the KT66. The R4, 7, 8, 9 all "float" on top
of the global NFB network. The feed from V1 anode to V2 grid is 6dB of positive FB.
You may think the distortion is increased 6dB as a result of the PFB. But the feed
allows V1 and V2 to be a balanced amp and most even numbered H are reduced.
Odd H may increase because of the PFB. In practice it is not a serious fault, because
output tube THD will always be much higher than drive amp THD even with the PFB.
Quad-II I have seen, they have not been serviced anyone
qualified and output
tubes have unbalanced anode currents and resistance values have changed and signals
to each output tube grids are badly unbalanced. THD can measure up to10 times more
than it should at all levels.
Curve A for original Quad-II without any Global Negative Feedback. To measure this,
the R10 100r is shunted to 0V, therefore not allowing the fed back voltage from Vo to
appear at V2 grid via R8 and at under R4 680r. The amp has 16r0 load connected to
OPT with 16r0 strapping, so this is class AB1 working and the first 9Watts are pure
One can say
that because the EF86 driver amp has low THD compared to output
then the measured THD without GNFB is mainly due to KT66. Notice that the THD goes
up to about 2.9% at clipping, and is 2% at 9Watts. It would worse if there was no CFB
winding which makes the KT66 act like they would if triode connected without CFB.
However, the CFB allows the higher Po max than when KT66 are triode connected.
Curve B is for the
same amp but with normal GNFB, ie, with no shunt across R10.
The average reduction of THD is about -24dB. At low levels, with 0.040Vrms input, the
Vo = 4.67Vrms with zero GNFB, and 0.48Vrms with GNFB, so the NFB gives about
-20dB gain reduction. The gain reduction with GNFB should be equal to THD reduction
and probably there is a less simply explained phenomena in the interaction of NFB
network and V1 & V2 tubes.
This is not
a bad set of measurements. THD spectra is mainly 3H at all
levels but has
many higher H present. At low levels, and when viewing the THD on oscilloscope,
its envelope is much modulated by rectifier noise. Some diode switching noise is present.
curves are for class AB1 Po, and have uneven curvatures between
high levels. This indicates there is some THD cancellation or addition between
input / driver stages and KT66.
the THD with 32 ohms with the 16r0 links and for pure class A
and with the GNFB the THD was below 0.1% at 20 Watts. What more could anyone want?
This is for a MkIV Dynaco Monobloc I recently re-engineered.
Now this is
a very cheeky sneaky move of mine. But whenever considering
we should always be prepared to compare one amp with another, lest we loose all
understanding of "better" or "worse."
THD curve is almost a straight line. That will be due to the
very low THD
input / driver I have in MkIV which uses 6CG7, which qualifies to be the King Of Little
reformed MkIV was tested with the same RLa-a as for Quad-II,
The output load is 4r0, less than the minimum 6r0 I recommend. The MkIV has 33%
UL taps with KT88, and 6CG7 input & drivers, and diode rectifier and B+ ripple noise
at OPT CT is -60dB. The MkIV has 14dB with 4r0 load, some 10dB less than Quad-II,
yet the Dynaco has THD at -6dB lower up to 2.2Watts. The Dynaco gives 38Watts
class AB1 max with 4r0, and only 5 Watts for pure class A. So while the Quad has
less than 0.1% THD at 9Watts, the Dynaco has 0.2% because it has moved into
class AB1 working. But the THD spectra in Dynaco THD is mainly 3H + slight 2H but
has much less other rubbish we see in Quad-II spectra. The majority of listening is
done with amp power less than 2Watts.
Dynaco's KT88 have Pda+Pg2 = 20Watts, so they run cooler than
the Quad KT66.
If the KT88 were idled with Pda = 30Watts, the first 9 Watts would have less THD than
Quad's. The class A Po is determined by RLa-a and the idle Iadc. KT88 can have the
same 70mAdc as KT66, but have higher Ea, and therefore will make more class AB Po
total. I doubt sometimes that "extra heat is worth the ears", but then my customers told
conclude that if one is to re-furbish such grand but limited old amps, it is always
possible to improve the circuit behavior to get a better measuring and better sounding amp.
Fig 2. Basic Reformed Quad-II
component numbers used here don't relate to any component
the original schematic except by coincidence. This schematic is a re-drawn version
I did in 2007, and includes slight changes.
improvements could be done but the above has what I consider to
minimum. Improvements include...
1, Remove old 2 pin mains Bulgin chassis socket to rubbish bin. Install IEC 3 pin
mains chassis plug with 2AG mains fuse included for standard IEC mains cable.
infuriate those wanting to keep the Quad-22 preamp arrangement
Safety comes first, and if you ignore this step 1, then don't blame me if you
electrocute yourself, or family member.
mains on-off DPST rocker switch. Bypass switches with 10nF 2kV
ceramics, ( not shown ).
RCA input socket just near existing Jones socket for Quad-22
If you doubt you'd ever spoil your listening with an original Quad-22 control unit,
then remove the Jones socket. Note C1A and R1A to deal with Vdc swings in
R1B between 0V at Jones plug and the chassis. This interrupts
earth loops which can cause hum with other audio gear connected, preamps,
CD players etc.
grey box with 16+16uF caps inside and put in rubbish bin.
Disconnect HT windings and 5Vac heater windings from GZ32 valve socket.
Decide if you really must have a dead tube in your amp to make it look right.
Install well thought out terminal strips to allow the B+ rectifier circuitry to be built.
Do not try to use GZ32 as a slow turn on series diode for B+ rail. I tried, and it
didn't work out.
component numbers do NOT relate to any numbers in original
Quad-II or any
other schematic except by coincidence.
HT is rectified with silicon diodes
through current limiting 47r to a100uF. B+ to OPT CT
is filtered with CRC using 100uF - 200r - 470uF, to get 100Hz ripple at OPT CT > 0.06V.
The original Quad-II choke is retained to filter the output tube Eg2 and B+ for input/driver
tubes. The intention for this design was to retain nearly original working conditions for the
output tubes which in this case were to be KT90 instead of KT66. The amp owner
already had an 8585 amp I made which used 4 x KT90 per channel, and he liked these
tubes. I found the KT90EH made in Russia to be extremely reliable.With the CRC filter
AND the silicon diodes, the B+ voltage drop across200r is quite tolerable, and the
+380Vdc is actually higher than in original Quad-II with struggling GZ32 with high
effective series resistance.
gave a lower amplifier output resistance than KT66, slightly
lower THD and
better maximum current ability to cope with the RLa-a load of 4k2 including high winding
resistances when 8r0 is connected and with OPT links set for 8r0.
KT90 cathodes are individually biased with R15 & C7 and R16
& C8 networks
of 470r / 10W and 1,000 uF / 63V respectively. Under dynamic music conditions the
very slow time constant of the bias circuit prevents much movement of Ek cathode bias.
Balancing of each output tube Ikdc bias is mostly automatic but in this pair of amps
I included use of a low -Vdc supply so that output tube grid voltage can be adjusted
from about 0V to -8V which is enough to get good Ikdc balance. See VR1, 10k0 pot,
linear, wire wound, and 3Watt rated.
condition of bias balance or any tube fault is monitored by the
2 LED which should
glow at equal brightness when Ik for each tube is within 5% of being equal. I did use
a pair of red LEDs in amps for 2006, but this schematic above shows YELLOW LED
for showing balance.
In 2014 I included the active protection circuit with
SCR and green LED for "amp on"
and red LED to indicate amp is turned off internally because of some fault because of
excessive Ikdc in output tubes.
balance is easy via a 6.3mm chassis hole and use of thumb nail
or small blade
to turn a screw head flush with chassis side. The screw is end of 6.3mm pot shaft with
a slot for a blade. GZ32 and the Jones 6 pin input socket have been removed to
rubbish bin. The 470uF B+ cap may be mounted where GZ32 and its socket used to be.
It is good practice to place a metal plate shield between 470uF cap and the output tube
standing nearby, so prevent the cap getting too hot from radiated heat from output tube,
which shortens its life.
heater winding for GZ32 is connected in series with one end of
heater winding which has a CT to 0V. Thus there is an available 8.2Vac voltage quite able
to drive voltage doubler circuits to produce + / 10Vdc. The + 10Vdc is CRC filtered for
6.3Vdc to V1, V2 heaters. The -10Vdc feeds the bias balance pot. V1 and V2 are ideally
EF80 / 6BX6, which are a "sharp cut off pentode" with 1,001 uses and which has about
50% higher gm than EF86 when compared using same Ia and Ea and Eg2. I have the
6BX6 set up as a long-tail pair differential amp and WITHOUT the PARAPHASE
connection which was used in the original Quad-II circuits. This means the odd number
HD of V1 is not fed to grid of V2, so the 2 tubes produce less THD. There is good
balancing of even number HD currents at commoned cathodes because the Rk is a
high enough value.
& V2 cathodes are supplied by a B- rail = -380Vdc produced
with diodes from
HT winding. The -380V is filtered with RCRC and a virtually constant current is supplied
to commoned cathodes of V1 & V2. I have used 3 series resistors ( R12 + R26 ) to
avoid excessive Vdc across any one R.
of R12 22k has been bootstrapped with 10uF to the GNFB applied
grid of V2. This makes R12 act like a resistance far greater than 22k. There will be
about 0.2Vac across 22k, so Iac = 0.009mA. Vk to 0V = 1.72Vac (max approx), so
the effective R looking down into R12 = 190k. The bootstrapping should make
V1 Va output no more than 2% above V2 Va output. For closer Va balance, use
about 4M7 across R16 330k.
Quad-II mods I have used a MJE340 as a CCS for commoned cathodes
The amp protection and bias balance circuit requires a small 5VA
240V : 12V
transformer, T3, mounted somewhere conveniently under the chassis. This provides
power for the small circuit board used for the solid state devices The picture image
at the top left of the page shows the amp balance LEDS with equal brightness.
there is 25 watts AB into 8 ohms with slightly more into 4 ohms,
Rout = 0.78 ohms. Some folks would hang me from an old oak tree after hotting
up a Quad II amp like I have. But I have thought about what happens in a fault
situation resulting from bias failure, shorted or saturated output tube, shorted
speaker cables, shorted coupling caps etc, etc.
bias balance LEDs have been adjusted for equal brightness, any
in output tube Iadc will always make one LED glow more brightly than the other,
which tells an owner to re-balance the bias. When the tubes continue aging and
becoming more unmatched, the owner runs out of turn on the pot and he cannot
adjust the pot for balanced bias currents and equal LED brightness. This tells
the owner to replace one or both output tubes. If the owner ignores the LEDs, then
Iadc in one output tube may rise into the dangerous region where a tube is too hot.
Many owners ignore such things and do not hear the degradation of music.
When a tube has too much Iadc, the Ek will rise enough to send a signal to the
SCR which turns off the amp automatically to save a huge expense on PT or OPT
etc. The owner cannot ignore this.
bigger tubes like KT88, KT90, or 6550 may overheat the
transformer primary winding and power transformer windings if one or both tubes
were to fail and become saturated with current resulting in a maximum of about
1Amp dc from the power supply. But such a failure would blow a slow 0.5A
mains fuse. I tried a slow blow 0.5A which worked OK instead of the original 2A fuse
shown on the original amp schematic. I've always thought the 2A fuse value was
too high for original Quads with 220V to 250V mains.
intermediate level fault is where ONE output tube conducts
before it melts down to become a pure short circuit between anode and cathode.
This condition of thermal runaway creates overheating in OPT and PT before an
output tube becomes a short, or goes open.
placing 1k0 from the OPT CT to 0V to simulate a serious but
fault event. This generated Idc = 350mA. A 0.5A slow-blow mains fuse blew after
a 3 second delay. But 2A fuse would not blow, so fuse could probably be 0.75A
fast blow, but a 0.5A slow offers sensitive protection.
suffer heating / cooling cycles at each turn on. The cycles
fatigue the wire
in fuse, so the more sensitive the fuse, the more you get "nuisance fuse blows" after
some weeks or months of use, so a compromise is to use a 0.75A slow fuse,
and never leave the amps turned on overnight, especially in hot weather.
output tube were to conduct say 200mAdc for say 5 minutes, then
will rise to +97Vdc. This may well cause the 1,000uF bypass cap rated for 63V
to become a short circuit and then Ikdc will increase to a maximum possible and
the mains fuse should blow soon enough. KT66 saturated current is about 300mAdc,
but KT90 may conduct 500mAdc. If there is a continual 500mAdc in 1/2 the primary
of Quad-II OPT which has Rw = 167r, then 41Watts of heat is generated in the
1/2 primary winding. Unless the amp is turned off well before Iadc reaches 500mA,
the OPT winding will fuse open well before you can feel the OPT getting HOT.
Even KT66 with max 300mA Idc will produce 15Watts of heat in 1/2 the primary
which will damage or fuse the winding.
It is no
good to use a fuse between cathode to CFB winding because it
to be 200mA rated to avoid nuisance blowing. If additional fuses are to be used,
place one 0.5A slow blows between each end of HT winding and subsequent
R and diodes. It is ALWAYS better to have an active protection circuit.
output terminals short circuited, and the input signal turned up
gross overload, a 0.5A mains fuse will not blow because the power supply anode
current in the seriously overloaded condition only increases by just over twice to
250mA which isn't enough to make such a low value mains fuse blow unless the
gross overload is sustained for minutes which is long enough for tubes to overheat,
get red hot anodes, then go into thermal runaway. Such a sustained fault like this
is very unlikely to occur because either silence from the shorted output or gross
distortion of sound woulds alert an owner that something is wrong and he/she
would turn things off and investigate speaker connections or speaker condition
If one of
two speakers is suspected of causing silence or high
reverse the speakers to opposite amp channels and if the problem follows the
change of channels then the speakers have a problem but amps are probably OK.
In the 2006
amps I didn't provide any means of automatic turn off of the B+
the whole amp if cathode Ek rose too high as in the Fig 2 schematic. This was
similar practice to a pair of amps I modded in 1999. I felt that the owner would
keep a fairly close eye on the bias balance LEDs which will tell him if something
is wrong. Well, that owner has Gone Upstairs and who knows who will use the
2014 schematic of this 2006 amp includes the active protection.
Since 2006, I have made circuit boards small enough to fit in any amplifier I ever
blow 0.5A fuse has proved to be quite reliable, and it survives
in-rush current at turn on. The slow blow fuse will tolerate short time input current
surges maybe up to 1.5A, but will blow when the AVERAGE current goes just
over 0.5 Amps and stays there for some time.
The Quad II set up as the above schematic draws 88Watts, so with 245V mains
the input current average is 0.36A. This is less than in the original Quad-II which
has the GZ32 heaters consuming about 12 Watts more.
KT90 are perfectly interchangeable with KT66 and draw the same idle current.
But KT90 can produce a outright maximum of 30Watts instead of 22 Watts
with KT66. KT90 heater current is 1.6A instead of KT66 at 1.3A but it is OK
because these amps were designed to have add on tubed preamps and tuners
which will never be used with this pair of amps. The musical performance includes
tighter bass and more controlled and detailed treble, so I have to say KT90 in
Quad-II sounds better. KT88 / 6550 can be used OK.
The right image has the whole system in trial use before sending it off to my
customer, although GZ32 are not plugged in to keep it looking original.
On the left
image you can see the IEC shielded mains socket that replaces
Bulgin original. The old Quad-II recessed 4 mm banana sockets were replaced
with something from some retired HP test gear because the originals had cracked
aging plastic and from years of speaker cables being wrenched sideways.
The new signal input socket is Canare 75 ohm RCA which replaces the 6 way
"Jones" socket used in the original amps. The terminals are mounted on a white
fibreglass panel. The appearance was not important; the SOUND and the
RELIABILITY were the important issues.
Shielded interconnect cables from a preamp should only be used since the speaker
output cables are close to the amp input. I did try using unshielded dual foil cables
which were close to the speaker cables. No HF oscillations occurred, probably
because the live input cable is tied to the low impedance of the cathode follower in
the preamp. But I do not like unshielded twisted pair or flat foil interconnects because
they often pick up switching noise and other noise from nearby mains cabling and
these cable types are fragile, and always break sooner or later.
I quite like RG58 coax cable for interconnects.
A very much
modified Quad 22 control units is to the left of the power amps
described in my page on Quad 22 mods. The separate preamp power supply is
on the shelf below the two amps. The 2 power amps and preamp each has a mains
turn on switch.
power amp I used a blue "on" LED and red mains on switch placed
in an aluminium panel to cover holes for original mains voltage settings, something
not needed since the 240V mains in Oz rarely varies, and is always over 240V
unless it is a cold night in winter. There is never any need to employ the other two
lower mains voltage settings. Blue LED are too bright with say 4mAdc, and they run
tolerably bright with about 0.25mAdc. Owners prefer blue, but I prefer green for
"on" using a plain diffused 5mm dia type which runs on 4mAdc.
who worry about THD, here is....
graph 4 is drawn
on logarithmic axis for both THD and output Power.
The test is for the Fig 3 REFORMED schematic of reformed Quad-II, and shows
results for KT66 and KT90 with same loading in the same circuit with same GNFB
which is much less than in original Quad-II amps.
However, in 2014 I tested an original Quad-II amp with
exact original schematic
and in fair condition, and the THD is plotted in Graph 3 above on this page. ))
In Graph 4
you can see that at onset of visible waveform flats on CRO, ie,
clipping, both KT66 and KT90 produce about 1% THD at 21Watts and 24Watts into 8
an average of 1/2 the THD produced by KT66, but make 1/3 of THD
3Watts which covers most listening levels.
KT90 make 0.03% at 3W, 0.1% at 14W.
KT66 make 0.1% at 3W, 0.2% at 14W.
The curves with all their kinks are typical class AB tube amp measurements when driven
with pentodes which produce more THD than triodes. But at clipping the THD and IMD does
not matter as much as at normal levels < 3Watts. This is where our focus on the soloist is
intense, but really, anything below 0.05% is not too bad, considering tube amp artifacts are
less objectionable than solid state's. The trend for the KT90 to have half the THD of the KT66
continues below 1/4Watt. In 2006, it was difficult for me to measure THD accurately because
at 1/4 Watt Vo = 1.41Vrms, and if THD = 0.01%, then the THD voltage is 0.141mV, and noise
can be easily 0.5mV which obscures observation on CRO of the THD. The noise becomes
the dominant artifact at low volume. If noise = 0.5mV, and does not increase at say 14Watts
where Vo = 10.6Vrms, then SNR is said to be -86.5 dB which is quite acceptable.
tried measuring THD with loads between 4 ohms and no load at
all. With a 16 ohm
load connected to the the amp set for 8 ohms, at 1Watt the THD is 6 dB less than with
8 ohms, and with 4 ohms its about twice what the 8 ohm produces. At low levels of below
2 Watts which covers much listening for many people, any load down to 4 ohms is OK, and
KT90 give around 1/2 the THD of KT66 for all loads.
measured output impedance with KT66 in my circuit is 1.2 ohms,
and with KT90 it is
0.9 ohms approx. I also tried Russian 6550EH which gave similar THD to the Russian KT90EH,
and Rout = 1.0 ohm. The other thing to bear in mind is that I have 15dB GNFB used in my
reformed Quad-II. In the original Quad-II with paraphase circuit for EF86, there is about 21dB
NFB, or 6dB more.
graph 3 above, and with KT66 in original amps, I got 0. 07% THD
In Graph 3 with KT90, and at 9Watts, I get 0.06%, and Rout is lower than original.
So, I get better THD results with 1/2 the applied GNFB.
The first Quad-II mods I did in
Fig 4. Schematic of the first
reformed Quad II I attempted in 1999, using KT88 in triode,
plus triode input tubes instead of the EF86 pentodes.
Fig 4 was hand drawn 3 years before I went online or had a PC or scanner. Part numbers
have no intentional resemblance to any other Quad-II schematics on this page. With mains
at 250Vac I found I had B+ = + 410Vdc after diodes into 100uF C18 with KT88 triodes as
There is no picture of the amp, but GZ32 and Quad-II choke was removed, and a new
aluminium box mounted above chassis and snug between OPT and PT and about 85mm
high. This box had ventilation holes to allow air flow from below chassis into box and out
of box side away from nearby output tubes. Inside the box I fitted a 2H anode choke
and other PSU caps. Fixed bias is used for about 55mA per KT88 connected as triodes.
original 20H Quad-II choke does not filter B+ anode supply for
KT66, and is used to
only filter KT66 screen supply, B+ for EF86 and B+ output to other components. With
original 16uF + 16uF caps, there is barely enough CLC hum filtering for B+ for KT66
screens and EF86 anodes. The anode supply in original amp was from first 16uF after
GZ32 and anode B+ filtering is quite unacceptable by today's standards.
better to have an anode filter choke which is at least as large
as original 20H choke,
but with Rw < 40r so wire size will be higher and turns much reduced, hence L will be lower.
The minimum L value should be 2H but can be 4H but it would be almost impossible to
make a choke of 4H with Rw = 40r, and with air gap to suit 150mAdc, and using wasteless
pattern E&I and no larger than the original Quad-II choke. However, a clever DIYer will allow
an anode choke to be taller than original, and will fit it inside a box with el-caps. And box
can be snugly between OPT and PT boxes above chassis and be up to same height,
same style of metalwork, but with ventilation, and painted to match the battleship grey
paint used by Quad, and which perhaps was once used in WW2 for camouflaging military
gear during Britain's terrible summer and winter weather.
The higher B+ voltage suits triode operation because of the
limitation of triode anode
voltage swing in class AB1 because of grid current.
schematic gave 20Watts in triode class AB1 and measures very
well. In 1999,
I used only 12dB GNFB which is much less than original Quad II. The amount of local
cathode FB in the output stage is not high because the triode gain is only half that for
tetrode-with-CFB. and fixed Eg2. The effective CFB is only 3dB. When trioded output
tubes are used, just enough local CFB exists to compensate for the high winding
resistance losses in the OPT. Better than nothing though.
CFB + GNFB applied was about 15 dB. I first used EL34 in triode with only
6dB GNFB but the owner said it gave poor dynamics when compared to the owner's
other 10Watt amp with 6GW8 output tubes in ultralinear with about 16dB of global NFB.
The use of KT88 in triode and 12dB GNFB delivered the kind of vibrant and accurate
dynamics he was looking for.
cannot use Quad II amps without global NFB because the output
will be too high even with triode wired output tubes. Rout was less than 1r0 as I have
it with 12dB GNFB and KT88 triodes.
Quad-II amps with KT66 without GNFB have and only local CFB have
Rout = 9
ohms measured when OPT is strapped for 9r0. With KT88 in triode, and without GNFB t
he Rout = 5 ohms and still too high for any speaker. The GNFB reduces the 5r0 to 1r0.
There is a
10k pot for balancing the idle Idc in each KT88. The value of
R21 is not
shown since it has to be trimmed to get 55mAdc per tube. The balance is set by
turning a pot shaft so each LED is equal brightness.
form the differential driver LTP amp and have a constant current
transistor in the commoned cathode "tail". The bjt's extremely high collector impedance
is above several Meg ohms, so it is impossible for the transistor to color the sound,
except to improve its quality, because it makes the 6CG7work with better balance and
less THD. The CCS ensures the pair of opposite phase output voltages will be exactly
equal even if the triodes are grossly unmatched, but only if each anode of LTP has
equal load resistances.
Input is a
12AT7, but could be 12AX7 or 12AU7 or 12AY7 with circuit mods to
anode and NFB resistors so that the gain and amount of NFB applied is appropriate.
This modification sounds very well even with 4 ohm speakers, but note that the OPT link
settings are for 9r0 on the terminals Q, R, S, T.
A pair of
amps with this mod has been used every day from 1999 to 2012 in a house
Cooma, NSW where summers are not cool. KT88 in Quad II amps don't cause
overheating unless the full range of add AM and FM tuners and Quad-22 control units are
used. KT88 use marginally more heater power than KT66. In these two amps I drilled
lots of holes in the bottom covers and reversed the position of plates holding the power
tube sockets lower than chassis top surface to get more air flow through the bottom of
the amp and up around the output tubes to help keep the amp cool.
Because there is no GZ32, and because KT88 draw less idle current than KT66 in original
amps, there is less heat generated in the power transformer and overall temperature
is always cooler than originals. I have not needed to repair anything in these amps.
And now I
have re-drawn the 1999 schematic in 2014 with most features
with some subtle changes which will definitely work better....
Fig 4A shows what I would do with trioded KT88 in Quad-II now. Notice the higher
B+ for V1 12AT7, and 6CG7 V2a + V2b with increase of Iadc, to give better dynamics.
The CCS is fed from a negative rail to maximize the Ea and to get a larger possible
Va swing at low THD. GNFB is increased 2dB. Just remember R9&C4, C11&R27,
R28&C13 are arranged for what I think will give unconditional HF stability with the
amount of GNFB which is derived from point Q, and not from speaker output terminal.
This schematic requires that the OPT anode and cathode connections are reversed to
get correct phase for GNFB. This is not shown clearly in my old 1999 schematic. Fig 4.
4B is the PSU for Fig 4A and contains more than I used in
1999, and all that is
relevant now is the better schematic without the shortcomings of the old Fig 4
schematic which did not contain automatic protection against excessive Ikdc in
one or both output tubes. Possibly there is no need for the Relay 2 which delays
the turn on for 4 seconds and limits inrush current, thus permitting a 0.5A mains fuse.
are two added 0.5A slow fuses for HT winding, needed in case
all after the fuses becomes a short circuit. Inside the original Quad-II PT the 6.3Vac
heater winding CT is connected to the HT CT, so just using a fuse between HT CT
and 0V is unwise because I have 1/2 the 6.3Vac winding seriesed with 5Vac winding
to give 8.3Vac from which +10Vdc and -20Vdc is derived as easily as possible for
V1 DC heating, and for +/- Vdc supply to differential amp with bjts to power the the
two yellow LEDs indicating Ikdc balance of KT88.
is reduced to -18Vdc with R15 68r and zener diodes to
stabilize the -Vdc
feeding CCS for V2 cathodes. The -75Vdc is developed with pair of 10uF x 450V
caps and RC filtering, and current is low so little heat is wasted in the series R
between -380Vdc and -70Vdc.
47nF C15, C16 shunt the ends of HT winding to 0V and
suppress switching noise
from diodes.The original Quad-II choke and GZ32 and tube socket are deleted.
I calculated a 3H choke with Rw < 40r plus two 220uF x 470V caps will give excellent
B+ anode supply filtering. Its not really necessary to use 470uF caps, my schematic
shows what will work OK and what easily fits in limited space.
220uF and all other B+ rail caps for V1, V2a, V2b anodes are
all rated for 450V
because B+ will all rise to +440Vdc after turn on but before tubes conduct. The caps
will fit easily into a ventilated box above chassis, with caps mounted on cool side of
box away from heat from KT88. The KT88 are not running any hotter than KT66 in
original Quad-II amps.
schematic is similar to Fig 3 above.
I have Q1 MJE340 for CCS to V1, V2 cathodes. I have Q2 & Q3, TIP31C, for
Dynamic Bias Stabilization for cathode bias. I included the pair of BJT in this amp
because it prevents any rise in Ek across C13, C14 due to rectification effects of signal
cathode currents of KT88 during class AB operation. The KT88 can generate more
maximum Ia than KT66, and the rise in Ek is greater. The BJTs still allow auto biasing
to occur which save having any bias adjustment to confuse & worry any owner.
I did not
include a bias balancing pot or bias balance LED indicators, but
minimize the active protection measures.
Over the years I found the individual cathode biasing will always give enough Idc
balance while ever tubes remain fairly serviceable. All that's really needed is active
protection to provide a "safety net" for our fallen angels, ie, something to turn off
the amp if a KT88 decides to conduct too much Idc, which most will do towards the
end of their life.
Fig 5 has
a table for with the same anode to
anode loading by OPT used for 3
different ways of linking OPT terminals Q to T, to get load matches for 16r, 9r and 4r.
tube loading between the two anodes may be calculated :-
RLa-a = ( OPT ZR x [ secondary load + secondary Rw ] ) + Rw whole primary.
In the 2010
amp I tested with 8r0 sec load with OPT links set for 9r0.
The OPT ZR = 432:1. Rw secondary = 1.08r. Rw primary = 334r.
RLa-a = (
432 x [ 8.0 + 1.08 ] ) + 334 = 3,456r + 466r + 344r = 4,256r.
The Rw %
losses are 100% x Rw total / Total RLa-a = 100% x ( 466 + 334 )
that if we measure 28.5Watts into 8r0, the tubes must make 35.1
Watts anode power, 6.6Watts of heating occurs to OPT windings. (( The OPT would
warm up with a constant 6.6Watts, but the duty cycle for music signal is low, so OPT
heating by AC Po is negligible, and primary wire heating from Idc is the only worry,
but this is usually less than 2Watts. ))
the amps produce enough clean power and without clipping for a
musical experience, winding losses have little significance to listeners. However, the high
Rw cheats the buying public of what could be much better with very little extra amp weight
or size. The high Rw makes OPTs fragile and prone to fusing from failing tubes.
The real winners from high winding losses are company accountants, always keen to
minimize quality and costs of production and to be sure of getting a handsome wage.
I have never ever employed an accountant.
Fig 6. PSU and Protection.
Fig 6 shows
essential features of a well made amp PSU while attempting to
as simple as possible. In the B+ rail, there is no CRC filter before the OPT anode
primary winding. This seems an error, but I found that there is not a big reason for CRC
as I have in other Quad-II amps because I wanted to keep the B+ as high as possible
and well regulated with minimum series resistance. The 100Hz hum at top of C17
470uF = 0.7Vrms, and is 1/25 that of an original Quad-II with a lousy 16uF used where I
have 470uF. But if any wishes to use an extra 220uF plus a 2H choke, then feel free, but
there is not much room under the chassis or above it.
there is a 0.5A fuse in series with all B+ anode Idc after C17
470uF. This is a
precaution against a sudden shorting of anode circuit to 0V. Such a short can cause the
energy stored in C17 to rapidly discharge through some part of OPT anode winding which
possibly may fuse the winding. Max Idc during a short could be 3Amps! It is a rare event
which I have never seen.
There is a delay circuit to limit inrush current for a few
seconds after each and every
time the amp s turned on. The inrush current is large without R15 because the HT
winding must charge directly into a large 470uF cap. Even when amp is turned off
with hot tubes, then on again within 3 seconds, the delay re-occurs and limits mains
peak current due to R15 270r until B+ is about 2/3 full value, and then R15 is shunted
by Relay 2. The inrush Iac when Relay 2 closes is no more than for initial turn on and
less than 1/2 what it is without the delay circuit. The delay circuit allows a sensitive low
value mains fuse without nuisance blowing.
6.3Vac for ALL tube heating in 2010 but in this schematic I have
used the 5Vac
heater winding plus 1/2 6.3Vac heater winding to get 8.2Vac for a voltage doubler to
make Vdc rails of approximately +/- 18Vdc. +18Vdc is RC filtered down to 12.6Vdc at
0.3Adc for 6BX6 heaters in series. The -18Vdc is RC filtered for the CCS for 6BX6
protection circuit has been simplified to what I found works
well. Instead of
having the cathode Ek at tops of R&C cathode bias networks divided and filtered and
then with diodes to gate of SCR, I have just used the winding resistance of the CFB
winding, at points U and W, to give a source of Vdc which can turn on the gate of
SCR. Each 1/2 of CFB winding has Rw = 17 ohms approx, so with 65mA, the Vdc
= 1.1Vdc. If this rises to 2Vdc, then it is enough to trigger SCR with 0.67Vdc at gate
and Ikdc cannot exceed about 120mAdc. If the cathode bypass caps were to ever
fail to a short, then the protection would work more reliably than if the sample Vdc
is taken from cathode at top of bypass caps. However, I have never ever witnessed
electrolytic cap failure unless the Vdc went too high for too long, or they became
so hot the liquid inside boiled. The SCR will be triggered well before electrolytic
caps ever get hot or short out.
amount of CFB + GNFB exceeds 20dB so there will be more tendency
oscillations. I have shown what I think may be required values for "critical damping", ie,
open loop gain and phase shift reduction above 20kHz by parts R10, C4, R26, C10,
R27, C11. The amp must not oscillate at any F above 20kHz even when loaded solely
by any pure C between 0.05uF and 2uF at low levels when open loop gain is highest.
The amp must be tested with a 5kHz square wave at low level, say 2Vpeak output and
with pure C load. The square waves should not have more than 6dB of overshoot, and
not more than say 4 ringing waves declining to a flat line within 100uS, or a 1/2 wave
time for 5kHz. If tested with sine waves with a C load, the response should not have
peaks in the response exceeding +6dB above the 1kHz levels.
Everyone building any tube amp MUST overcome the tendency of all amps to oscillate
when ANY NFB is applied. The amp is a bandpass device with NFB applied around the
open loop gain and there are limits to how much NFB can be applied and how much
bandwidth is possible when NFB is applied. Everyone needs to understand phase shift basics.
one, I have HT winding charging 220uF via 1N5408 without current
It may seem odd, but in fact the PT has enough Rw to prevent peak charge current being
too high during normal operation. The CLC filter with 220uF, 3H, 220uF have been chosen
because they smooth the ripple well enough and size of parts are kept low. There is no
need to use the old Quad-II 20H choke, and we need the room for 3H choke.
The PSU as
it is may be used for KT88 in triode if the screens are
connected to anode
instead of to the B+ supply at OPT CT.
the possible use of having each KT88 screen fed by 2k2 from OPT
each bypassed with100uF to cathode of opposite KT88. This is a way of applying a
10% Ultralinear connection which increases the local screen CFB between g2 and k
from 10% to 20%. It slightly reduces KT88 gain and Ra and THD. Whether it is worth
the extra pair of R&C may be argued.
But I have
always found class A1 operation of pentodes or tetrodes to sound
and measure best when both adequate CFB is used in conjunction with some screen
signal derived from UL tap in anode winding. In this case there are no available UL
taps on Quad OPT, unless one alters the OPT, and then its not much worth the
trouble when some screen signal with wanted phase can be derived from cathode of
opposite KT88. One could also just bypass screens to cathode of the same KT88,
and then you have the KT88 acting as a pure tetrode with only CFB applied between
g1 and k. This means the worse sounding H spectra in THD of pure tetrode remain,
although are reduced with the CFB. The effective UL% changes H spectra favourably
before the CFB is even applied.
At turn on,
there is a 4second delay before R17 270r is shunted by Relay 2,
delay is long enough for B+ to rise to near +350Vdc and the Mains Iac peak is kept low.
The circuit board needed for small parts for delay, protection, and balance indication
will need to be kept as small as possible, and made with care, and installed with screw
fixing off spacers, and after removing 2 screws, the board should fold out on flexible
wiring to allow access to parts.
Output Transformer issues for Quad-II amps.
This picture from Keith Snook's website.
Figures 9,10,11,12 show what is inside the Quad-II OPT can.
Fig 12 shows a small alteration to OPT secondary winding connections sealed inside pot
containing the OPT to allow all sec windings to be better used for use with 4r0 speakers.
There must be one added turret terminal to allow the alteration, and it involves removing
the OPT from its can. A 2mm brass bolt with a nut could be used instead of a turret.
Extreme care must be used for this modification, because inexperienced fools could so
easily ruin a good OPT.
possible is the creation of OPT UL taps from anode primary turns
joins between N1-2 to N2-3, and N6-7 to N7-8.
I don't know what this R is for, or what the ohm value is. Possibly it helps
equalize current density in Sec windings and helps amp stability. Peter Walker had
a reason for each and everything to be found or not to be found in Quad-II amps.
For a better OPT, a
much larger core and completely different design would be used for
better overall performance while being easier to wind, with less than half the high
original resistance which can fuse open so easily.
If anyone can find someone to custom build OPTs, here are a few suggestions:-
Fig 13. Small size OPT...
This OPT has a core plan size which cannot fit inside the original can, but the OPT
should fit on chassis to replace the can position. The windings will project beneath
the can position and whatever is in the way would need to be moved.
A bell end top cover would look silly, but a sheet steel lid made to appear like the
top of original can could be screw fixed to angles clamping E&I lams.
The transformer and lid can be painted to match Quad's Battleship Grey.
Fig 14. Large size OPT
Fig 13 is the biggest practical OPT for Quad-II. The core plan sizes are 96mm x 80mm
and this will not fit inside original can. The original amp must be totally re-engineered.
The EF86 sockets and circuit board are removed. The screen supply filter choke and
GZ32 socket are removed. 1mm metal plates are needed under the new OPT to
fully cover holes for EF86 sockets and to suit re-installation of 9-pin sockets where
rectifier and choke have been removed. A metal cover over replacement OPT should
be made to suit the shape of power transformer can.
Now to explain some benefits gained.
The original Quad-II OPT has well interleaved windings with 7 primary sections
and 6 secondary sections. This gives good HF response, and yet the amp can oscillate
at HF. The cause is probably due to EF86 pentodes having very high Ra and high RLa
values so that a small amount of shunt C causes enough phase shift to become a
problem above 20kHz. Quad-II will benefit much by using critical damping Zobel
R&C networks and the use of a trioded EF86 input and twin triode 6CG7 or 12AU7
LTP as driver. The electronic dynamics are optimized and and intermodulation products
of input/driver are reduced, and this translates to better sonic purity.
The OPT LF performance is not as good as many other OPTs. The original OPT
is only 1/3 the weight it really should be. Just consider the Williamson OPT which used
E&I laminated core with of 32mm Tongue x 44mm Stack of "Super Silcor" E&I lams
with non wasteless pattern and window size of 75mm x 25mm.
Everyone laughed at Mr Williamson, but Mr Walker probably didn't, because he would
have known the Williamson design was better, but for commercial reasons Walker used
a toy sized OPT. Electronic commercialism meant prices for amps of any quality never
had prices which were toy-like. But in UK in 1950, costs of all materials were high, and
recovery after WW2 was slow.
Williamson OPT details are in RDH4, page 748, half way down.
For best bass performance the Fsat of OPT should be below 20Hz for the rated
power Vac applied across the OPT winding. RDH has some wise comments which
infuriate all accountants and bosses of any companies making amplifiers in 1950s.
Many think the primary inductance is the most important LF parameter for bass F.
It is not. It is the distortion free performance at full Po down to a low F, ie, down
to 20Hz which is necessary, and if attained, automatically leads to enough primary
inductance. If you test a tube amp using a pink noise source test signal with randomly
varying F and amplitude from dc to 30kHz, you will see how inadequate many
tube amps are because there's constant saturation of core and instant huge
intermittent bursts of high IMD each time some low level 4Hz wave saturates the
Most music does not have much signal below 10Hz, but we live in a world with
deep bass and "sub-woofer" signals in music, or movie sound tracks so the amps
must be "ready for anything". Core saturation is independent of loading or signal
current and is a "voltage caused" phenomena, with magnetic field about proportional
to the applied Vac and F.
The older core material had a low Bac max, maybe 1Tesla
for poor E&I lams, but even with modern GOSS it is not wise to go more than 1.2T
when harmonic distortion, mainly 3H very rapidly exceeds 3%. at some F below 100Hz.
So if the core saturates at say Va-a = 400Vrms at 40Hz, Bac = 1.2T, it means that
20Hz the B = 1.2T at 200Vrms, and 10Hz the Fsat occurs at only 100Vrms, a very
low level of Po. Therefore all tube amps should have C+R input high pass filtering to
exclude F below 10Hz, and have open-loop gain and NFB arranged to give sharp cut
off below 5Hz, and have an OPT with low Fsat < 20Hz at full Po rated Va-a.
Nearly every tube amp manufacturer has tried to dodge the issues of quality determined
by best engineering principles.
Fsat can be calculated = 22.6 x Vrms x 10,000 / ( Afe x Np x 1.2Tesla )
where Vrms is across OPT primary, 22.6 and 10,000 are constants, Afe is square mm,
Np is primary turns, and 1.2 is the maximum magnetic field strength, Bac.
For original Quad-II, maximum Va-a is in the class A condition with RLa-a about 9k0,
with Ea about = +350V, and Ia = 70mAdc in each KT66.
Va = 317Vpk, 224Vrms, so Va-a max = 448Vrms giving 22Watts into 9k0. Winding
losses reduce this to about 19Watts at output terminals.
For determining Fsat, winding losses are ignored because the Fsat is considered where
tubes are not loaded with RLa-a, but VaPk is equal to maximum Va-a for class A,
or about 0.9 x Ea.
So Quad-II Va-a max = 448Vrms. Afe = 25mm x 25mm = 625 sq.mm,
and Np = 3,180 turns.
Fsat = 22.6 x 448 x 10,000 / ( 625 x 3,180 x 1.2 ) = 42Hz.
For a lower Fsat, Afe must be increased, or turns increased. But the primary turns
already have excessive winding resistance which can lead to a fused OPT.
Increasing Afe is the easiest solution, but that means turn length increases and
higher winding resistance. So we will have to reduce Np.
Consider small 40W OPT, Fig 13, when used in identical conditions.
Fsat = 22.6 x 448 x 10,000 / ( 28 x 80 x 1,500 x 1.2 ) = 25.1Hz. The primary wire
copper section area is 1.6 times larger, and the Np less than 1/2 of original, and
Rwp will be less than 1/2 the original 334r, and winding loss% much less.
There is less interleaving in 40W OPT, but with less turns the Lp inductance remains
quite high enough. The leakage inductance is also proportional to Np squared, so with
1/2 the Np of original OPT the LL reduces to 0.25 x original for the original 7P x 6S
interleave pattern. In practice the 40W OPT has similar low LL to original with
interleave pattern = 5P x 4S. The shunt C is low enough.
If you disagree, then feel free to use my pages on OPT design to compare the figures
Consider larger 45W OPT, Fig 14.
Fsat = 22.6 x 480 x 10,000 / ( 32 x 75 x 1,820 x 1.2 ) = 19.2Hz, splendid, IMHO.
Consider the Williamson, from 1950,
Fsat = 22.6 x 480 x 10,000 / ( 32 x 44 x 4,400 x 1.2 ) = 13.6Hz. This seems better
than almost anything we see made now. The Williamson OPT was meant for 10k
RLa-a load, and 480V makes 23Watts into 10k0. The use of 4,400 primary turns
on Afe = 32 x 44, 1,408sq.mm means the Lp max is huge, ensuring LF stability with
20dB NFB and without Zobel networks and open loop gain shelving networks.
But in fact the Lp at low 6.3Vrms Va-a levels was barely 100H where core µ is low
because the permeability µ and inductance varies with applied Vac. But the high Np
also meant LL is high, and W saw the need for 5P x 4S interleaving. Remarkably,
the W OPT had good HF properties.
In today's world, we have cheap GOSS cores and cheap wire, and with a larger
core than used in 1950 the Np can become lower for the same outcome, and the
larger wire size is easier to wind. Williamson Afe = 1,408sq.mm, more than twice
Afe for Quad-II, My above 45W OPT has Afe = 2,400sq.mm, some 3.8 x Quad Afe.
Rwp for 45W OPT = 138r, and if load is 6k6 then P loss% = 2%, way below Quad-II.
The secondary losses = 2.2%. Total winding losses = 4.2%. The 448Va-a into 6k6
produces 30Watts AB1, with a high amount of class A. Use of KT88 will allow higher
Ea = 380V, and with RLa-a = 2k5, ( minimum rated RLa-a ), Va-a = 367Vrms to make
54Watts at anodes, with lower Sec RL of 1r5. Winding loss% increases as RL is
reduced, and is 11.1%, so expect a maximum of 48Watts class AB1 max. The KT88
will allow load mis-matches without clouds of smoke, and a 4r0 speaker with dips in Z
to 2r0 will not make music sound bad.
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