Continued from SE OPT Calc Page 2,
SE OPT calc page 3 Contents :-

Practical Testing of 8 Watt SEUL OPT for 1 x EL34 for old AM radio.
Fig 19. Schematic for rather good 8 Watt SE amp.
Notes about schematic etc.
Fig 20. Photo of old radio after restoration.
Notes on testing amp and OPT.

Oscilloscope pictures of waveforms produced with 8 Watt SEUL OPT.
CRO 1. Healthy wave at clipping power at 1kHz.
CRO 2. Measuring Lp.
How to adjust the air gap. 
Fig 21. Wasteless pattern lamination dimensions.
ALTERNATIVE METHOD to calculate max PO of any pentode
CRO 3, 4, 5, Waves during core saturation phenomena.
Fig 22. Graph of Air gap Vs Fsat and Lp.    
CRO 6, 7, 8, Waves at -6dB level for 47Hz, 20Hz, 16Hz.
Conclusions about SEUL for 1 x EL34,
Notes and calculation checks for all.

SE OPT Easy Method for calculating any SE OPT.
For Basic Parameters only.
1E.  Calculate RLa and PO.
2E.  Calculate Afe.
3E.  Calculate required Lp.
4E.  Calculate Np.
5E.  Calculate minimum primary wire size.
6E.  Calculate minimum core window size L x H.
7E.  Calculate Core T + S.
8E.  Calculate µe.
9E.  Check Lp, Fsat.
10E. Calculate Air gap.

High Voltage testing of transformers.
Fig 23. Schematic for testing insulation of transformer.
OPT3 bobbin details for 25W OPT from web pages created May 2006.

OPT3 from 2006.
Fig 24. Details of OPT3.


Practical testing of 8 Watt SE Amp example, measuring Lp,
and Fsat, and setting the air gap.

TUBE AMP VOLTAGES !!!!!!!!!!!!!!!

Fig 19. 
Fig 19 resembles a very good type of 8W audio amplifier
for an old radio where there may have been a 6V6 used to make 3 watts.
Usually the best old AM radios will have a B+ supply generated from a
HT winding with a 5Y3 rectifier giving about +280Vdc available
at up to 50mAdc at the SE OPT connection. In many old AM radio sets there
may be a field coil choke on the speaker with Rw = 1k5, and then there is a large
voltage drop across the 5Y3 anode resistance and across OPT winding resistance.
Electrolytic filter caps are often quite low so ripple voltage is high.

The ancient 1940 speaker may be in such poor condition it is best to replace it.
Usually the existing OPT will need to be replaced because of its poor bandwidth,
high winding losses, and incorrect impedance ratio. A modern replacement speaker
with no field coil may have lower sensitivity but have a better bass response.
Treble response in an old speaker may have only extended to 3kHz, so a small
coned HF unit may be mounted concentrically inside the bass unit to extend
HF to give almost hi-fi performance, with at least 9kHz BW.

To drive a modern speaker, the audio amp may be vastly improved. The 5Y3
is replaced with silicon diodes charging into a CRC filter using modern
replacement 220uF x 450Vdc rated caps and R may be say 235 ohms using a
pair of 10W rated 470 ohms in parallel. Usually 70mA may be had from an
existing old power transformer which may be still serviceable, and not on the
verge of burning out with repeated continued use. Some power transformers
will benefit from being removed from the chassis, and soaked in electrical
varnish over night, then baked for 4 hours at 125C the next day. 

With the reductions of series resistance in the PSU, the Vdc at C1 at 70mA
dc draw may be  may be +420Vdc, and at the OPT connection at C2 is +400Vdc.
Such a nice high B+ allows the use of an EL34, 6L6, ( even 807 ) to be used in
the chassis socket meant for 5Y3, and where the 6V6 once was, there may be
a 12AX7, or 6J7 in triode mode. I find the 12AX7 plus EL34 in triode mode
is simplest and will give 6 watts even if using the old OPT if it is suitable.
Fig 13 above shows a best practice use of OPT with SE+CFB configuration.

If there is a moderate Vdc drop across PSU resistance and OPT primary and
across the cathode biasing RC network then often it is possible to get a
healthy Ea across the EL34 of +350Vdc, and up to 8 Watts of output power.
The OPT shown in Fig 13 has 15% CFB windings, but a simpler effective OPT
may be made with one anode winding with a CT which is used for a 50% UL tap
to the EL34 screen.

The new OPT with larger core size and lower winding losses will give little
B+ voltage drop but the EL34 could have Ea at 300V and and still produce
better power and lower THD than a 6V6, 6F6 etc or 6BQ4/EL84.
Triode operation is also generally superb compared to normal 6V6/EL84
in pentode mode, and with triode connected EL34 there is no need for global
NFB. Often it is found the old ancient OPT and speaker made prior to 1955
(when accountants took over design sections of radio production) are OK and
will work very well with EL34 in triode. Triode operation of EL34 is best
with Ea at +350V to +420V, and Ia at idle should be adjusted to give the
most power with least THD at 400Hz. Often the Ia will be no more than what
was used in the original use of 6V6, so there may be no extra heat produced
in the power transformer. The 6L6 or 807 beam tetrodes need more grid drive
signal, but draw less screen current, but sound excellent in triode or with 50%
SEUL or with 15% CFB.
I have recently re-built a few ancient radios from the 1935 to 1950 period
using SEUL or triode connected EL34. I have quite a few to use, and when
set up with Pda = 18Watts, they will last maybe 15 years easily. I used a
second hand EL34 in the AM radio I designed and built in 1999, and it
produces excellent sound and has never needed to be serviced.
My radio customers appreciate me putting a pair of RCA sockets which may be
used for CD, FM tuner, whatever, for very pleasing mono sound for intimate
lounge room or kitchen listening.

Fig 20.

Fig 20 shows the rear view of a completely re-engineered 1935 AM radio chassis
from July 2011. The modern 10" bass-midrange speaker with concentric HF driver
is shown. There is a mains fuse, IEC mains cable input and grounded chassis for
safety. There is a replacement PT on rhs, with EL34 and 12AX7 audio amp.
There is a new OPT in the chassis centre. There are RCA terminals provided
for use with FM tuner or CD player. Sound will be mono, but very listenable.
The other tubes used to replace rare European odd ball types from 1935 are
6AN7 mixer, 6N8 IF amp, 12AU7 detector, 12AU7 treble control stage with
slight gain.

The OPT has a non-wasteless pattern of old low grade EI laminations
with Tongue = 25mm, Stack = 24mm, Window L = 52mm, H = 16.6mm.
ML = 189mm. Np = 3,200 turns of 0.3mm Cu dia wire.
Primary nominal RL = 6,400 ohms.
Secondary windings may be configured to give 4//81t for 4.1ohms,
3//108t for 7.3 ohms, 2//162t for 16.4 ohms, but the 7.3 ohm sec is used
because the speaker for this project is 8 ohms.

The OPT in Fig 14&15 was calculated for use in an old floor standing AM radio
where low power operation will be used.
The 10" speaker is mounted in
a large open backed cabinet which has a natural cut off at about 60Hz
There is no point in making an amp which needs to produce full power at
below 50Hz. The amplifier bandwidth has been tailored to have a cut off at
Fsat at 0dB output level could be allowed to be higher than for a true
hi-fi amplifier with Fsat and cut off at below 20Hz. 

Notes about testing the OPT and amp performance :-

Before  proceeding to test the SE audio amp the circuit needs to be checked
3 times against the schematic before turning anything on. Fuses should be
used in mains supply and between HT winding CT and 0V, so that you
don't ruin an old power or output transformer or a field coil choke or anything
else. Typical HT fuse might be 0.5A fast blow, or a value which will offer
protection if there is bias failure in the output tube when Ia may rise to 300mA
before the output tube self destructs. The mains fuse should be 240V x 0.5A
or 110V x 1A slow blow, or whatever value is needed to prevent nuisance
fuse blowings, but never so high to offer almost no protection.

Once the winding of the OPT is complete, the UN-VARNISHED OR
UN-WAXED OPT is temporarily placed into the amplifier which is
fully operational with all NFB
loops connected.

If the bobbin has been varnished with cold cure polyurethane two pack
mix, the core may be assembled into the wound bobbin without the final
application of applied varnish to the completed item which is best done
by potting in a 50-50 mix of dry clean dry sand and slow setting epoxy
casting resin.

The core air gap MUST be able to be adjusted before the final potting
or varnishing procedure, because after potting it is impossible to easily
change the air gap.

The air gap used initially should be what has been calculated,
and it may
or may not need any adjustment.

The amplifier output voltage at clipping should be measured using a dummy
resistance load which is the load value for absolute maximum possible output
power using a sine wave, with THD < 2%, and at 400 to 1,000Hz.
With higher output voltage levels the sine wave should show equal clipping to
positive and negative wave crests, ie, the wave clipping is symmetrical.

The output voltage at just under 2% THD at the onset of symmetrical
wave clipping at 1kHz with
a sine wave at 1kHz is the 0dB reference
voltage level.

An oscilloscope must be used to monitor all signal voltage measurements
and distortions at the output terminals. Sine waves with THD < 0.5% must
be used to drive the amp from a signal generator capable of F range between
2Hz and 2MHz and with a flat response for all ranges which may be adjusted
for level and F.

To monitor all the anode currents, 10 ohms resistance is placed in series at
anode to OPT, as R15, Test Point 2. The schematic also shows 10ohms below
CFB winding as R14, giving Test Point 1 so monitoring anode signal current is
more convenient than having the R at a high voltage level. The R14 also carries
the screen dc current which will be about 7% of anode current. There is some
screen signal current flow so measurements of anode current are most accurate
at TP2. But anode current waveform distortion is most easily seen at TP1.
With GNFB present, the OP voltage at Sec may appear linear while the anode
current may appear slightly distorted because the NFB is keeping the Sec voltage
linear by applying a grid signal voltage containing an "error signal" to the output
tube grid. As the output signal is increased towards clipping the output tube
grid signal and anode current signal will show increasing wave distortions, but at
about 2dB below clipping, all wave distortions should be less than 2% and thus
distortion will not affect measurements and calculations. 

The amplifier should be able to be optimized for unconditional stability at LF
and HF so that with GNFB connected it will not oscillate at any F even with
no secondary load connected. If no GNFB is to be used, the EL34 should be
strapped as a triode. Slightly less output power is available than pentode mode
but there will less distortion and measurements will be easier because the Ra
is so much lower than with a 6V6 or 6BQ5 in tetrode or pentode. EL34 in
triode gives Ra = 1k3 approx, which may be 1/5 of RLa, so damping factor
with triode is high enough to not need GNFB. With 50% SEUL, Ra will be
about 3k0, and some GNFB is useful, but seldom is there any need for more
than 12dB of GNFB. I do not like ordinary pentode connection which requires
more GNFB. 15% CFB will reduce EL34 Ra to less than triode. 

Below are photos of oscilloscope screen using a 1983 dual trace 15Mhz
Hitachi oscilloscope ( CRO. )

The images below show typical wave forms common to many tube
SE amps
with Global Negative Feedback, GNFB.
The amp tested to gain these images has 1 x Sovtek EL34 set up in SEUL
configuration with UL tap at 50% of anode turns.

CRO 1.
CRO 1 shows two traces, top is the secondary output voltage
with rated
load at onset of symmetrical clipping at 1khz.
The negative wave crests at Ea minimum show clipping because of grid
current and the inability of the EL34 anode voltage to swing any lower.
The positive wave crests at Ea maximum are just about to reach Ia cut off.

The bottom trace shows Ia signal wave across R14 in the Fig 13 schematic.
Peaks on current wave are produced by effect of NFB trying to drive tube
into producing more current and force Ea to swing lower.
The Ia is monitored at the cathode and the current peaks produced by
grid current and screen current peaking because the load voltage shows
no such current peaks. At this point the coupling cap driving the EL34
is gaining a negative charge from grid current which tends to make
the tube effectively biased to conduct less idle Idc.
Sustained over drive slightly upsets DC working conditions. 

The 0.0dB Reference VO signal would be set slightly below levels shown
which indicate THD = 2% approximately.

Measuring primary inductance.
The primary inductance should be measured at the -6dB Vo level
at the lowest frequency easily measured and where THD < 3%.

CRO 2.
CRO 2 top trace shows Vo = -6dB, 25Hz, NO RL. At this voltage level
there is no saturation and the primary load is solely the inductive reactance.

The bottom trace shows the anode current signal flow which is seen
across R14 in Fig 13 schematic.
There is no severe distortion and EL34 load at the primary is a linear
inductive reactance without saturation effects.

The anode to cathode Vac is measured and in this case = 100Vrms.
This is the total Vac signal across all primary windings.
If 15% CFB windings are used, there would be 100Vrms measured
between anode and cathode, with 15Vrms between cathode and 0V,
and 85Vrms between anode and 0V, with the two voltages Va and Vk
having opposite phases. But with SEUL, there is just 100Vrms between
anode and 0V, or across the whole anode winding from B+ to anode.

In this example Vac across R14 = 0.265Vrms, and so Ia = 0.0265Arms,
So Lp reactance, XLp, at 25Hz = Vac / Iac = 100V / 0.0265A = 3,773 ohms.

Lp = XLp / ( F x 6.28 )
where L is Henrys, F is in Hertz, and 6.28 = constant = 2 x pye.
So Lp = 3,773 / ( 25 x 6.28 ) = 24H. 

Because XLp = 3.8k approx, the Vo wave shows about 2% THD because
of the loading effect of a reactance approaching 1/2 the ideal RLa value.
The bottom 25Hz current waveform shows 5% THD because the GNFB
is trying to correct the voltage distortion. If you examined the EL34 grid
signal you might find it had 5% THD or more, because a fraction of any
THD at the amp output is fed back to V1 12AX7 cathode then amplified
so it appears at V2 EL34 grid with its phase inverted, therefore trying to
reduce the output THD as it is produced.

The above waveforms show that the air gap would be nearly optimised.
If there was severe THD at say -6dB at say 60Hz, it may indicate the gap
is far too small or too large or that the OPT being tested may be unsuitable
for the intended application.

The frequency response for where THD < 2% may be plotted on a
graph at 0.0dB, -6dB, and -12dB. -6dB is where Vo = 1/2 Vreference,
and -12dB is where Vo = 1/4 Vreference. Graphs for F response may
be done using an exercise book page and pencil and oscilloscope used to
measure relative voltages.

There are blank sheets at the bottom of this page which you may print out
to enable graphs to be made in your workshop with a pencil. 

How to adjust the air gap size for optimum operation.

Fig 21.

Fig 21 shows wasteless pattern E&I laminations.
There are TWO magnetic paths around the TWO core windows which act
together as ONE magnetic path.
The Iron magnetic path length, ML = 2H + 2L + 22H/7 for any core material.
Wasteless Pattern E&I lams, ML = 5.57 x T.

The calculated gap is initially used for tests if it was calculated.
If the gap was not calculated or preset, which may be the case with an old
stock OPT being trialed, then use at least 0.05mm gapping material.

The dc flow in the OPT core will usually be enough to draw the blocks of
Es and Is tightly together and leaving the clamping bolts loose may allow
the E&I to come as close as the gap material permits. But a newly wound
OPT may be found to be slightly forcing the stack of Es and Is apart slightly,
so mechanical clamping together may be necessary. This is a real danger to
the OPT performance and MUST be checked very carefully during tests.
Hand made bobbins using say 1mm thick cardboard are very prone to
slight bulge during winding and and inexperienced DIYers may not design
the bobbin size with some clearance, or may not have a winding lathe which
clamps the bobbin properly.

The OPT may tend to howl while testing with applied sine waves because
of some slight movement of windings and core before final varnishing.
Without any DC flow in the core, the core may be easily prized apart to
enable enough layers of paper to be carefully inserted so they lay flat and
make up close to the calculated air gap. Once the calculated paper gapping
has been inserted, the yoke bolts are very slightly tightened and the Is tapped
tp to be tight against the Es. With C-cores, the clamps around the cores are
slightly drawn up.

Adjustments to the gap size can only be done with the B+ turned off so Idc
does not flow thus permitting the E&I to be prized apart to remove or add
sheets of gap material during observations of primary inductance and Fsat.

Adjusting the air gap.

The signal voltage at Sec should be set at the 0.0dB reference level
at 1kHz with the rated load, and measured. The load is then removed,
and output voltage will rise so it should be then adjusted down to equal
the reference 0.0dB level. The distortion without any load should be
lower than with a load. 

The frequency is then reduced slowly down and it should be able to be
reduced to at least 100Hz without any serious increase in distortion.

But as F is reduced below 100Hz, some THD will inevitably become
visible on the CRO screen.

CRO 3.
CRO 3 shows top trace has Vo = 0.0dB, 100Hz, With NO RL.
The primary inductance is the only load, and in this example the Lp was
found to be 24H measured at the -6dB level at 25 Hz. At 100Hz, Lp
reactance of the Lp > 15k
. The Vo wave has low THD because of the
high RLa value and because the EL34 has slightly more gain with the
high load thus making the effective applied GNFB = 13.5dB approx.
The bottom trace shows Ia current waveform has some distortion
because the iron is beginning to show some non linear behaviour.

CRO 4.

CRO 4 top trace has Vo = 0.0dB, at 32Hz, with NO RL. The primary
reactance has begun to become a non linear reactance suffering partial
magnetic field collapse during part of each wave cycle. The actual Fsat
onset was at approximately 35Hz.
The Vo wave has a sudden onset of
the high THD seen above. 
The bottom trace shows the 32Hz current waveform with high distortion.
The iron has become highly non linear, and all music would be ruined.

CRO 5..
CRO 5 top trace has Vo = 0dB, 25Hz, NO RL. The primary reactance
has become a very non linear reactance suffering substantial magnetic field
collapse during each wave cycle.

The bottom 25Hz current waveform contains extreme distortion. The iron
has become highly non linear. 

So from these observations we see Fsat = 35Hz. Now the air gap material
thickness should be increased by using a sheet of paper at 0.05mm and the
Fsat re-measured by observing the frequency of onset of saturation
distortion. The inductance must also be re-measured, and may be done at
the Fsat onset, but at the -6dB Vo level.

I fill one exercize book each fortnight with notes and schematic drawings
related to my work and to whatever else I think about. Each book has 120
sheets, and is 6mm thick, so each sheet = 6mm  / 120 thick = 0.05mm.

The use of paper is a convenient way to adjust an air gap, but if you have
some polyester sheeting of known thickness it is also quite suitable.

When the gap is reduced slightly, the Lp inductance may increase.
If Lp increases, but Fsat may occur at too high a frequency. 

The Fsat should be found to be a lower F as the air gap is increased.

If the LP inductance reduces with increased air gap, its value must be
recorded for the loading effect it may cause.

Fig 22.

Fig 22 has three graph curves for the SE OPT used for the example amp
shown in Fig 13 which has EL34 for CFB operation. The SE OPT could
be used for SEUL if all the primary turns are used in the anode circuit
and there are suitable screen taps.

The air gap initially calculated and used was 0.25mm, with gap material
of approx 0.12mm thick.

This gave Fsat = 35Hz, and XLp = RLa at 43Hz. These F seem high,
but in many old radios you would find Fsat = 80Hz and XLp = RLa at 100Hz.
The designers intended for low power levels and bass response extending
down to only 120Hz. Hence the inadequate walnut sized OPTs which were
finally approved by the company accountant who dines with the CEO.

If the air gap is reduced to a minimum, ie, with no gapping material,
one might expect the maximum iron µ to be 2,500 which occurs if the E&I
lams are fully interleaved. But with butted stacks of Es and Is, the core
acts as though there still is a gap, but in fact, there isn't. The change of grain
in crystalline structure and the imperfections of the butted join will limit µe
to much less than if all E&I were maximally interleaved which might give µ
max of 2,500, and where µ max is as low as 2,500, then the maximum µe
with butted Es and Is may be less than 750. The Fig 17 graph for Lp becomes
uncertain below where the gap < 0.15mm.

So, reducing the gap to the minimum might increase Lp to 38H, and thus
give XLp = RLa at 30Hz, and raise Fsat to 60Hz. Having such a high Fsat
is more undesirable than having negligible loading with high Lp reactance
at 60Hz.

If the gap is increased above 0.25mm to say 0.35mm, then Fsat moves down
to 28Hz, and XLp = 18H, with XLp = RLa at 55Hz. This will mean the tube
will see a partially reactive load = 0.707 x RLa, ie, 4k5 at 55Hz, and thus
tube distortion at 0dB will begin at F above Fsat. This is preferable to having
Fsat occurring above F where load reduces. 
In other words, if Fsat is low enough at 0dB, and XLp may be less than RLa
at a higher F, and it is a bonus if XLp = RLa at a lower F than Fsat.

If the gap is further increased to say 0.45mm, then Fsat moves only slightly
lower to 26Hz, while Lp becomes 15H and XLp = RLa at 70Hz which I
think is too high, even for a humble AM radio OPT.

In this case, from the Fig 17 graph we may read off the ONE frequency
where XLp = RLa AND where Fsat occurs at 0dB output level.

This is seen where the Fsat B curve intersects the XLp C curve at point P
which is at 40Hz.
The air gap required to achieve this F may be read vertically below P on the
air gap axis and = 0.22mm.
Adjusting the air gap so finely may not be easily possible if there is not any
sufficiently fine enough material to use.

If the air gap was left at 0.25mm then Fsat = 35Hz with XLp = RLa at 43 Hz,
then the operation is quite OK for this application.

This example OPT for a radio OPT will be found to sound quite superb at
ordinary listening levels, providing the source signal has THD < 1% at say
4Vrms from the AM detector, or 1Vrms from CD player, or FM tuner.
Normal audio detector circuits used in 99% of old radios fail to produce
low distortion.

For a true hi-fi OPT for use with a single EL34 to make 8 Watts,
then one would begin with AFe = 450 x sq.root PO.

In this case max PO will be 0.45 x tube pda.
If Pda = 18Watts then PO = 8.1Watts.
So Afe = 450 x sq.rt 8.1 = 1,, and the core should have
Tongue = 32mm and Stack = 40mm. This core size is obviously much larger
than is ever seen in most
SE high-end hi-fi amps producing 8 watts from
tubes like 6550, KT88, EL34, 6L6 or 300B, but to obey all the rules
mentioned at this website, one must start with much more iron than in most
amplifiers made in large batches with design work by Bean Kounter, that
utterly despicable character who seems to find employment to remove
quality everywhere he goes.

What I have described here is acceptable for the intended use and it
demonstrates how to set air gaps to maintain a high enough ohm value
for the anode load down to low enough bass frequencies and without
core saturation for most music.

CRO 6.
CRO 6 has Vo = -6dB, F = 47Hz and with secondary RL so PO = 1.85 Watts.
There is no sign of saturation or loading related distortions.
The bottom Ia trace is also very clean.

Within music signals, the bass between 50Hz and 200Hz may average 4 times
levels above 200Hz, which means that one would not want bass signals to exceed
the -6dB level so there is voltage headroom for midrange&treble levels without
much clipping. An ancient old radio made in 1937 with large floor standing cabinet
may have a 10" speaker driver with sensitivity at 1kHz = of 93dB/W at one metre,
and only give a minimal amount of bass. The OP tube might be a 42, 6F6, 6V6,
6BQ5, 6M5 and because OPT winding losses might be 25%, and B+ rather low,
PO max might only be 2.5 Watts, so at the highest practical level of -6dB Vo,
PO = 0.63 Watts, which is enough to fill a lounge room with sound.
But so often the old 1935 speaker has so many faults that it must be replaced
with a modern 10" driver unit which has a flatter response curve and much higher
and more pleasing bass level relative to its midrange than the 1935 driver.
But the modern speaker sensitivity may be 90dB/W/M at 1kHz, and people
expect less THD/IMD and higher levels, so it is prudent to upgrade the maximum
possible PO to 8 Watts or more so that healthy average levels can be achieved.

Most modern drivers with permanent magnets and more supple cone suspension
diaphragms will operate with far less THD or cone break up than anything made
in 1935. HF performance of all 10" drivers is usually poor with cut off at 2kHz, and
I recommend that a 65mm cone drive unit be used to cover the response between
2kHz and 8kHz which is sufficient for excellent radio sound for most listeners.
In a recent example, I made a holding strap for the small driver using an aluminium
section of 20mm x 3mm so the smaller driver could be mounted concentrically
"inside" the larger bass unit. This meant there was no need to RUIN a beautiful old
timber cabinet by trying to make a 75mm hole for an additional HF driver.

Some drivers meant for use in car audio systems or for ceiling mounting have
a small cone midrange and dome tweeter already concentrically mounted.
These make good replacements for many 8" 1935 speakers where the original
speaker was also 8" ( 200mm dia ) The speaker must be obtained before the
OPT is wound to ensure the OPT has the right turn ratio and impedance ratio.

CRO 7.

CRO 7 top trace shows Vo = -6dB, F = 20Hz, with NO RL. The load
on the EL34 output tube is a mainly linear inductive reactance of 24H.
The reactance of 24H at 20Hz = 3k0, less than 1/2 the rated load of 6k4,
so hence one can see approximately 3% THD caused by the low value
inductive load at LF.
The bottom trace for Ia shows THD at about 7
%. The loop of GNFB
is not succeeding to reduce the voltage distortion very much. But because
little music content occurs at 20Hz, the sound will remain OK.

CRO 8.
CRO 8 top trace has Vo = -6dB, F = 16Hz, with NO RL. The load on the
EL34 output tube has become a non linear inductive reactance of 24H
and saturation is occurring. In fact the onset of saturation at -6dB is about 17Hz,
and half the Fsat for the 0dB Vo level. This shows shows saturation to be a voltage
caused phenomena with no relation to the RL load used. The EL34 is struggling
with the low value inductive load which at 16Hz = 2k4. If the Vo was inspected
at the -12dB level, and at 8Hz, RL would be 1k2, and saturation would have just
The bottom trace for Ia shows THD at about 15
%. The loop of GNFB
is not succeeding to reduce the THD at Vo.

Conclusions for this example for 1 x EL34, SEUL.
Fsat should be found using Vo = 0dB.
This 0dB voltage level is obtained at the symmetrical clipping level at 1kHz
with the RLa load value which gives the maximum possible SE output power.

The Fsat is measured at the 0dB level without any RL connected and where
saturation distortion becomes visible on the CRO, ie, exceeds 2%.

For pentode and tetrode output tubes without any NFB, the LF response of
the amplifier without any RL connected will reduce at approximately 6dB
per octave below where XLp = Ra. If Lp = 24H, and EL34 Ra = 12k0, then
Fco = 80Hz. It should be possible to have GNFB connected to maintain 0dB
Vo level below the unloaded Fco until saturation occurs, or where XLp = RLa
ie, the loaded Fco, and if RLa = 6k4, loaded Fco = 42Hz.

Fsat at 0dB level should be below the loaded Fco frequency. Where there is
no load used, the Fsat can usually be measured at the 0dB level at the loaded

The XLp may become too low at LF if Vo is maintained at the 0dB level,
and THD will begin to exceed 2%. The response should be plotted for Vo
levels limited by THD = 2%, and the Fsat may be found at the lower Vo
level than at 0dB.

Often such a situation may suggest the air gap is too small, and it may be
increased to lower the loaded Fco while raising the Fsat, until loaded Fco
= Fsat and both are at satisfactory minimum frequency where the level
is at 0dB.

Measurements of inductance should be made at between 20Hz and 50Hz
and Vo levels below which saturation or inductive loading distortion becomes
clearly visible on the oscilloscope.

In the 8 Watt SEUL OPT example, the Lp was found to be
24.2H at 32 Hz.

At what F is Lp reactance = RLa?
RLa = 6k4, and F = RLa / ( 6.28 x Lp )
= 42.1Hz.

Is this acceptable? In previous calculations with example OPT4,
The design principles try to give XL = RLa at Fsat = 14Hz for a good 40
Watt hi-fi amplifier.
But the example here is for only 1 x EL34 used in SEUL mode to work
in an AM radio to give about 3 times the 2.5Watts from a 6V6 coupled to
an ancient and very inefficient and inferior quality OPT.

Fsat at 42Hz is OK for an old radio. 

From measuring Lp, the iron µe is calculated :-
µe =  1,000,000,000 x Lp x mL
        1.26 x Np squared x S x T
= 1,000,000,000 x 24.2 x 189 / ( 1.26 x 3,200 x 3,200 x 24 x 25 )
= 590.

Bdc is then calculated :-
Bdc = 12.6 x µe x Np x Idc
              mL x 10,000
= 12.6 x 590 x 3,200 x 0.051 / ( 189 x 10,000 )


Bac is then calculated at the onset of saturation Fsat :-

Fsat = 35Hz, Va = 220Vrms at 0dB at 7.6W into 6k4. 

Bac   =    22.6 x V x 10,000 
                  S x T x Np x F

= 22.6 x 220 x 10,000 / ( 25 x 24 x 3,200 x 35 )
= 0.74Tesla.

Total Bac + Bdc = 0.64 + 0.74 = 1.38Tesla.

Is the air gap correct?

Calculate air gap using estimate of maximum permeability for the example,
perhaps 3,000 :-

Air gap = mL x ( µ - µe )
                   µ x µe

= 189 x ( 3,000 - 590 ) / ( 3,000 x 590 )
= 0.26mm. Use 0.12mm gap material across the two magnetic gaps.

What are consequences if gap is changed?

Fig 17 graph above shows results gained with changes in air gap.

Increase air gap to lower µe to say 300.

Lp will become 12.3 Henrys.
XLp = RLa at Fco and will become 82Hz.

The Bdc will reduce to 0.38Tesla.

Allowable Bac = 1.38 - 0.38 = 1.0 Tesla, which means higher Vac could
be applied at 35Hz without saturation, but Vo could not reach 0dB level
because XLp is too low.

Therefore air gap must not be increased.

Decrease air gap to obtain µe = 750 which probably would be the maximum
possible µe with normal air gapping without partial air gapping.

Lp will become 30.8 Henrys, and XLp = RLa at Fco = 33Hz.

The Bdc will increase to 0.94 Tesla.

Allowable Bac = 1.38T - 0.94T = 0.44Tesla, which means Fsat will occur
at a higher F.

Fsat  =  22.6 x V x 10,000 
             S x T x Np x Bac

= 22.6 x 220 x 10,000 / ( 25 x 24 x 3,200 x 0.44 )
= 59Hz.

This is too high and therefore air gap should not be reduced to increase µe.

All things considered, the air gap for SEUL operation is about correct.

Triode operation instead of SEUL could possibly be used instead of SEUL and PO
max could be about 4.4Watts with EL34 Pda = 17.8W. Load line analysis
would have to be done.

Triode operation may require higher Ea and lower Ia to gain the Ea swing and also
higher RLa but the Fco and Fsat and Lp must all be examined lest music be spoiled
by not optimizing the design.
Some may find this an easier method for SE OPT......

1E. Calculate RLa, maximum PO.

For SE tube or paralleled tubes the Center Value RLa is first calculated from
Eadc and Iadc. The Center Value RLa is the load where maximum possible
SE power is possible for the given Ea and Ia. Other loads need not be considered.

Pentodes, Beam Tetrodes, UL, CFB RLa = 0.9 x Ea/Ia,
Max PO = 0.45 x Ea x Ia,
Also Max PO = 0.5 x RLa x Ia squared.

Triodes, RLa = (Ea/Ia) - (2 x Ra),
Max PO =
0.5 x RLa x Ia squared.

eg, For 1 x EL34, SEUL, Ea = 350V, Ia = 50mA.

RLa = 0.9 x 350 / 0.05 = 6,300 ohms.

PO = 0.5 x 6,300 x 0.05 x 0.05 = 7.87 Watts.

2E. Calculate Afe,

Calculate Afe = 450 x sq.rt PO.

Eg, For SEUL EL34 pentode Afe = 450 x sq.rt 7.87 = 1,

3E. Calculate required Lp so XLp = RLa at 20Hz.

Lp for hi-fi = RLa / ( 6.28 x F ) and for 20Hz,
Lp = RLa / 125
eg, For 1 x EL34, SEUL, RLa = 6k3, Lp = 6,300 / 125 = 50.4H.

4E. Calculating primary turns, Np.

Np = Lp x Idc x 1,400,000

eg, For 1 x EL34 SEUL example,
Np = 51.2 x 0.05 x 1,400,000 / 1,262 = 2,839 turns.

NOTE, the above simple calculation was derived as follows :-

p = 1.26 x Np x Np x Afe x µe  .........equation 1    
           1,000,000,000 x ML

For cores with Idc flow in windings in one direction,
µe = Bdc x 10,000 x ML ......................equation 2
         12.6 x Np x Idc

Bdc will remain constant.
Bac + Bdc at Fsat at 20Hz, should not be more than 1.4Tesla to allow for
lowest grade iron which may be available from surplus re-cycled cores.
Higher grade GOSS cores will have a magnetic headroom for 1.6Tesla,
but the 1.4 Tesla design limit allows for slight variations.
Therefore Bdc for all designs should be 0.7Tesla.

µe = 0.7 x 10,000 x ML  = 555 x ML ......................equation 3
         12.6 x Np x Idc
       Np x Idc 

Therefore, this µe from eqtn 2 can be substituted into eqtn 1 to give 
Lp = 1.26 x Np x Np x Afe x 555 x ML .........equation 1    
           1,000,000,000 x ML x Np x Idc

Lp =     Np x Afe       
1,430,000 x Idc
Then Np = Lp x Idc x 1,400,000

The constant of 1,430,000 may be rounded to 1,400,000.

5E. Calculate minimum size of primary wire.

Calculate expected maximum Idc in any part of primary winding.
Idc = anode Ia dc plus screen Ia dc = Ia + Ig2 = approximately 1.25 x Ia.

Minimum Cu wire dia = 0.8 x sq.rt Idc at idle.

DC current density must not exceed max = 2A per

eg, For 1 x EL34, Ia = 50mA, Iga = 12mA, allow Idc max = 62mA.
Cu dia minimum = 0.8 x sq.rt 0.062Amps = 0.199mm.

Therefore minimum size primary wire = 0.20mm Cu dia minimum.

From wire table, oa wire dia = 0.245mm.

6E. Calculate minimum winding window size area
and L x H dimensions.

H = 1.092 x sq rt Np x oa dia.

eg, For 1 x EL34, H = 1.092 x sq rt 2,839 x 0.245mm = 14.25mm.
Therefore L = 3H = 43.5mm.

This is derived from :-
Most cores have window dimensions with aspect ratio 3:1, or L = 3H,
so area of window = 3H x H = 3 x Hsquared.
The primary wire occupies window area = 0.28 x L x H = 0.28 x 3Hsquared,
= 0.84 x H squared.
The primary wire occupies window area = Np x oa dia squared.
So 0.84 x H squared = Np x oa dia squared.
H = square root of ( Np x oa dia squared / 0.84 )
= sq.rt Np  x oa dia / 0.916

7E. Calculate Core Tongue and Stack, T and S.

The minimum window H dimension has been calculated above.

Consider wasteless pattern E&I laminations.
T = 2H.

eg, For 1 x EL34, from Step 6E, T = 2 x 14.25 = 28.5mm.

Select from the range of standard size T for old wasteless E&I lams,
0.75" = 19mm;  1.0" = 25.4mm;  1.125" = 28.57mm;  1.25" = 31.75mm;
1.5" = 38.1mm;  1.75" = 44.45mm;  2.0" = 50.8mm;  2.5" = 63.5mm.

The standard wasteless T could be 28.57mm ( 1.125" ).

Stack S = Afe / T = 1,262 / 28.57 = 44.17mm ( 1.74" ).

Consider non wasteless pattern E&I or C-cores.

H and L and Afe all must not be smaller than calculated so far.

Therefore choose the C-cores or non standard E&I lams based on the
window size.
For E&I, measure the T dimension and Stack = Afe / measured T.
Some samples of non standard wasteless E&I lams have T size less
than 2H. For example, if H = 14.4mm, T might be 22mm.
So if Afe = 1,, S = 1,262 / 22 = 57.4mm.

For C-cores, the S dimension is the "strip width" of the wound core
material. Measure the build up thickness of one C-core,
and Strip width = Afe / ( 2 x build-up ).
The build-up of a C-core is the height of the wound core material
which has been wound with many layers and
glued together.
Double C-cores are usually used hence T = 2 x build-up.

For example, go to the Eilor website which lists C-core sizes,
Consider the window sizes available and select the cores with nearest
window size *above* that which has been calculated so far.
eg, For 1 x EL34 SEUL, H x L = 14.4mm x 43mm, and the Eilor C-core
has a "T32" C-core with the following dimensions :-
Window = 15.9mm x 50.8mm.
Build up = 10mm,
Strip width = 32mm.
Therefore T = 20mm, so required strip width = 1,262 / (2 x 10) = 63.1mm.
Therefore one might use 4 x T32 cores, ie, two pairs of 00 double C-cores 
stacked on top of each other to give an Afe section = 20mm x 64mm.
The benefit of the larger window allows the wire size to be considerably
increased to reduce all winding losses. The Eilor C-cores are very nice
things to use, and are beautifully made, but the price is many times ordinary
wasteless pattern E&I lams. A very fine 8 Watt SE OPT may be made
using some old re-cycled core material which may not have cost anything,
except the labour involved to extract the material from old transformers
which have burnt out windings.  

The secondary pattern and accurate fitting of wire, insulation must
be worked out using the complex longer methods above.
8E. Calculate µe, assume wasteless E&I laminations.

µe = 3,111 x T / ( Np x Idc )

eg, For 1 x EL34, the core is wasteless, 28mm tongue size, Ia = 62mA
µe = 3,111 x 28 / ( 2,839 x 0.062 ) = 495.

This has been derived as follows:-
Assume always Bdc = 0.7T.
For wasteless pattern E&I, ML = 5.6 x T,
µe = Bdc x 10,000 x iron ML
12.6 x Np x Idc
= 0.7 x 10,000 x 5.6 x T / ( 12.6 x Np x Idc )

If µe < 750, µe may be achieved by normal air gap adjustment.

Now maximum possible Bdc
= 0.7 Tesla = 12.6 x µe x Np x Idc / ( 10,000 x ML ).

For a given RLa, and the same PO, Idc must not be changed.

If µe was calculated > 750, then to achieve lower µe the Np must be
increased, or ML made longer, ie, use a larger core size, say T = 32mm
with ML = 180mm. For a given Afe, the µe must not be reduced by
increasing the air gap because the Lp would then become too low.
With most E&I, even if the Es and Is are tight together without
an actual gap, µe may not rise above 750 unless the material is GOSS.
Partial air gapping might be used but it involves more patience and trial
and error assembly and testing.
But with C-cores the µe may easily be varied from anywhere between
max µ of say 10,000 to 100.

9E. Check Lp and Fsat calculations...
Lp = 1.26 x Np x Np x Afe x µe   
           1,000,000,000 x ML

eg, For 1 x EL34, Np = 2,839t, Afe = 28mm x 44mm, µe = 495,

Lp = 1.26 x 2.839 x 2.839 x 28 x 44 x 495 / ( 1,000 x 5.6 x 28 ) = 39.5Henrys.

If Lp is not as high as wanted in Step 3E, Core S may be increased
easily because Lp is proportional to S.

eg, For 1 x EL34, For wasteless E&I, the standard plastic bobbin size available
might be for 51mm stack or perhaps 57mm so S = 50mm gives Lp = 44.8H,
and S = 57 gives Lp = 51H which will be enough.

Check Fsat.
At Fsat, Bdc = Bac = 0.7Tesla,
Fsat = 22.6 x Va x 10,000  
Afe x Np x Bac

PO = 7.87 W into 6k3, Va = 223Vrms, Bac = 0.7Tesla,
Fsat = 22.6 x 223 x 10,000 / ( 28 x 44 x 2,839 x 0.7 ) = 20.6Hz OK.

Notice Fsat is inversely proportional to Afe, or S, so using higher S
means lower Fsat.

10E. Calculate Air Gap and gap material thickness.

Ag = ML x ( µ -µe )
             µ x µe
eg, for T = 28, ML = 5.6 x T = 157mm, µe = 601 from Step 6E.
Assume E&I core has max possible µ = 3,000.

Ag = 157 x ( 3,000 - 601 ) / ( 3,000 x 601 ) = 0.2088mm.

Therefore the gap on two sides of magnetic path = Ag / 2 = 0.1044mm,
say 0.1mm.


The OPT should survive HV testing without the OPT having been varnished
or waxed, and should survive the application of +4,000 Vdc to the primary
for 1 minute without any arcing with all the secondaries and core well grounded.

Fig 23.

Fig 23 shows a schematic for applying approximately +3,920Vdc to the
primary of a transformer under test via 9 x 1M 2W metal film resistors.
The rectifier is built with a "ladder network" to step up the 200Vac from
an old surplus transformer secondary using 0.1uF caps rated for 630V
and 1N4007 diodes.
If there is an old radio set transformer with secondary of 380V-0-380V,
the available Vac max is 760Vac, or 1,070V peak available. Therefore
the ladder network would need only 4 step ups to generate approximately
4,000Vdc, but the 0.1uF caps used should be 2,000V rated and the each
diode "rung" in the ladder would need to consist of say 4 x 1N4007 in series
each with 1Meg across each to ensure the reverse bias across each diode
is limited to less than 300Vpk, and about equal for each diode. 

The +HVdc may be varied by using a variac to control the primary of the
supply transformer.

The core and secondary is connected to a 10k0 x 5Watt resistance which
is taken 0V of the power supply which MUST be also grounded directly
to the green and yellow Earth wire from the wall power outlet. A Vdc
voltmeter is connected across the 10k0 resistance. 

When power is turned on and voltage is raised slowly to maximum, the
volt meter should not show any voltage across the 10k resistance. But if
an arc does occur, there will be about +4,000Vdc across 9 Megohms and
Idc = 0.44A. This current will flow through the 10k0 and produce a reading
of 4.4Vdc. This is the maximum current flow, and indicates a short circuit
somewhere between primary and anything at earth potential. If arcs occur,
they may be intermittent and pulses will be seen on the meter. If a constant
Vdc is seen across the 10k0, it indicates there is some DC current leakage
through what must be resistance in the insulation, and insulation is faulty.

No damage or smoke should occur to anything during testing.

The meter used for the measurement may be a normal cheap analog type.
Arcs may pulse, and digital meters may not give read out numbers which
can be understood.


Output transformer OPT3 for a previous design
detailed in 2006 edition of the website.
OPT No3 is capable of around 25 Watts of SE power output.

Fig 24.
Schematic, OPT
          No1, SE mode.

Readers may wish to analyze what tubes and operating conditions may
be suitable for OPT3.

Back to SE OPT calc Page 2.

Back to SE OPT calc Page 1.